Load control device for high-efficiency loads

ABSTRACT

A two-wire load control device (such as, a dimmer switch) is operable to control the amount of power delivered from an AC power source to an electrical load (such as, a high-efficiency lighting load) and has substantially no minimum load requirement. The dimmer switch includes a bidirectional semiconductor switch, which is operable to be rendered conductive each half-cycle and to remain conductive independent of the magnitude of a load current conducted through semiconductor switch. The dimmer switch comprises a control circuit that conducts a control current through the load in order to generate a gate drive signal for rendering the bidirectional semiconductor switch conductive and non-conductive each half-cycle. The control circuit may provide a constant gate drive to the bidirectional semiconductor switch after the bidirectional semiconductor switch is rendered conductive each half-cycle. The bidirectional semiconductor switch may comprise, for example, a triac or two field-effect transistors coupled in anti-series connection.

CROSS REFERENCE TO RELATED APPLICATIONS

This is a continuation of U.S. patent application Ser. No. 14/083,968,filed Nov. 19, 2013 entitled LOAD CONTROL DEVICE FOR HIGH-EFFICIENCYLOADS which is a continuation of U.S. patent application Ser. No.12/952,920, filed Nov. 23, 2010 by Robert C. Newman, Jr. et al. now U.S.Pat. No. 8,664,881 issued Mar. 4, 2014 entitled TWO-WIRE DIMMER SWITCHFOR LOW-POWER LOADS which is a non-provisional application of U.S.Provisional Patent Application No. 61/264,528, filed Nov. 25, 2009, andU.S. Provisional Patent Application No. 61/333,050, filed May 10, 2010,both entitled TWO-WIRE ANALOG DIMMER SWITCH FOR LOW-POWER LOADS, theentire disclosures of which are hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to load control devices for controllingthe amount of power delivered to an electrical load, and moreparticularly, to a two-wire analog dimmer switch for controlling theintensity of a low-power lighting load, such as a light-emitting diode(LED) light source having an LED driver circuit or a fluorescent lamphaving an electronic dimming ballast.

2. Description of the Related Art

Prior art two-wire dimmer switches are coupled in series electricalconnection between an alternating-current (AC) power source and alighting load for controlling the amount of power delivered from the ACpower source to the lighting load. A two-wire wall-mounted dimmer switchis adapted to be mounted to a standard electrical wallbox and comprisestwo load terminals: a hot terminal adapted to be coupled to the hot sideof the AC power source and a dimmed hot terminal adapted to be coupledto the lighting load. In other words, the two-wire dimmer switch doesnot require a connection to the neutral side of the AC power source(i.e., the load control device is a “two-wire” device). Prior art“three-way” dimmer switches may be used in three-way lighting systemsand comprise at least three load terminals, but do not require aconnection to the neutral side of the AC power source.

The dimmer switch typically comprises a bidirectional semiconductorswitch, e.g., a thryristor (such as a triac) or two field-effecttransistors (FETs) in anti-series connection. The bidirectionalsemiconductor switch is coupled in series between the AC power sourceand the load and is controlled to be conductive and non-conductive forportions of a half-cycle of the AC power source to thus control theamount of power delivered to the electrical load. Generally, dimmerswitches use either a forward phase-control dimming technique or areverse phase-control dimming technique in order to control when thebidirectional semiconductor switch is rendered conductive andnon-conductive to thus control the power delivered to the load. Thedimmer switch may comprise a toggle actuator for turning the lightingload on and off and an intensity adjustment actuator for adjusting theintensity of the lighting load. Examples of prior art dimmer switchesare described in greater detail is commonly-assigned U.S. Pat. No.5,248,919, issued Sep. 29, 1993, entitled LIGHTING CONTROL DEVICE; U.S.Pat. No. 6,969,959, issued Nov. 29, 2005, entitled ELECTRONIC CONTROLSYSTEMS AND METHODS; and U.S. Pat. No. 7,687,940, issued Mar. 30, 2010,entitled DIMMER SWITCH FOR USE WITH LIGHTING CIRCUITS HAVING THREE-WAYSWITCHES, the entire disclosures of which are hereby incorporated byreference.

With forward phase-control dimming, the bidirectional semiconductorswitch is rendered conductive at some point within each AC line voltagehalf-cycle and remains conductive until approximately the next voltagezero-crossing, such that the bidirectional semiconductor switch isconductive for a conduction time each half-cycle. A zero-crossing isdefined as the time at which the AC line voltage transitions frompositive to negative polarity, or from negative to positive polarity, atthe beginning of each half-cycle. Forward phase-control dimming is oftenused to control energy delivered to a resistive or inductive load, whichmay include, for example, an incandescent lamp or a magnetic low-voltagetransformer. The bidirectional semiconductor switch of a forwardphase-control dimmer switch is typically implemented as a thyristor,such as a triac or two silicon-controlled rectifiers (SCRs) coupled inanti-parallel connection, since a thyristor becomes non-conductive whenthe magnitude of the current conducted through the thyristor decreasesto approximately zero amps.

Many forward phase-control dimmers include analog control circuits (suchas timing circuits) for controlling when the thyristor is renderedconductive each half-cycle of the AC power source. The analog controlcircuit typically comprises a potentiometer, which may be adjusted inresponse to a user input provided from, for example, a linear slidercontrol or a rotary knob in order to control the amount of powerdelivered to the lighting load. The analog control circuit is typicallycoupled in parallel with the thyristor and conducts a small timingcurrent through the lighting load when the thyristor is non-conductive.The magnitude of the timing current is small enough such that thecontrolled lighting load is not illuminated to a level that isperceptible to the human eye when the lighting load is off.

Thyristors are typically characterized by a rated latching current and arated holding current, and comprise two main terminals and a controlterminal. The current conducted through the main terminals of thethyristor must exceed the latching current for the thyristor to becomefully conductive. In addition, the current conducted through the mainterminals of the thyristor must remain above the holding current for thethyristor to remain in full conduction. Since an incandescent lamp is aresistive lighting load, a typical forward phase-control dimmer switchis operable to conduct enough current through the incandescent lamp toexceed the rated latching and holding currents of the thyristor if theimpedance of the incandescent lamp is low enough. Therefore, prior artforward phase-control dimmer switches are typically rated to operateappropriately with lighting loads having a power rating above a minimumpower rating (e.g., approximately 40 W) to guarantee that the thyristorwill be able to latch and remained latched when dimming the lightingload.

With reverse phase-control dimming, the bidirectional semiconductorswitch is rendered conductive at the zero-crossing of the AC linevoltage and rendered non-conductive at some point within each half-cycleof the AC line voltage, such that the bidirectional semiconductor switchis conductive for a conduction time each half-cycle. Reversephase-control dimming is often used to control energy to a capacitiveload, which may include, for example, an electronic low-voltagetransformer. Since the bidirectional semiconductor switch must berendered conductive at the beginning of the half-cycle, and must be ableto be rendered non-conductive within the half-cycle, reversephase-control dimming requires that the dimmer switch have two FETs inanti-serial connection, or the like. A FET is operable to be renderedconductive and to remain conductive independent of the magnitude of thecurrent conducted through the FET. In other words, a FET is not limitedby a rated latching or holding current as is a thyristor. However, priorart reverse phase-control dimmer switches have either required neutralconnections and/or advanced control circuits (such as microprocessors)for controlling the operation of the FETs. In order to power amicroprocessor, the dimmer switch must also comprise a power supply,which is typically coupled in parallel with the FETs. These advancedcontrol circuits and power supplies add to the cost of prior artFET-based reverse phase-control dimmer switches (as compared to analogforward phase-control dimmer switches).

Further, in order to properly charge, the power supply of such atwo-wire dimmer switch must develop an amount of voltage across thepower supply and must conduct a charging current from the AC powersource through the electrical load, in many instances even when thelighting load is off. If the power rating of the lighting load is toolow, the charging current conducted by the power supply through thelighting load may be great enough to cause the lighting load toilluminate to a level that is perceptible to the human eye when thelighting load is off. Therefore, prior art FET-based reversephase-control dimmer switches are typically rated to operateappropriately with lighting loads having a power rating above a minimumpower rating to guarantee that the lighting load does not illuminate toa level that is perceptible to the human eye due to the power supplycurrent when the lighting load is off. Some prior art load controldevices, have included power supplies that only develop small voltagesand draw small currents when charging, such that the minimum powerrating of a controlling lighting load may be as low as 10 W. An exampleof such a power supply is described in greater detail incommonly-assigned U.S. patent application Ser. No. 12/751,324, filedMar. 31, 2010, entitled SMART ELECTRONIC SWITCH FOR LOW-POWER LOADS, theentire disclosure of which is hereby incorporated by reference.

Nevertheless, it is desirable to be able to control the amount of powerto electrical loads having power rating lower than those able to becontrolled by the prior art forward and reverse phase-control dimmerswitches. In order to save energy, high-efficiency lighting loads, suchas, for example, compact fluorescent lamps (CFLs) and light-emittingdiode (LED) light sources, are being used in place of or as replacementsfor conventional incandescent or halogen lamps. High-efficiency lightsources typically consume less power and provide longer operationallives as compared to incandescent and halogen lamps. In order toilluminate properly, a load regulation device (e.g., such as anelectronic dimming ballast or an LED driver) must be coupled between theAC power source and the respective high-efficiency light source (i.e.,the compact fluorescent lamp or the LED light source) for regulating thepower supplied to the high-efficiency light source.

A dimmer switch controlling a high-efficiency light source may becoupled in series between the AC power source and the load controldevice for the high-efficiency light source. Some high-efficiencylighting loads are integrally housed with the load regulation devices ina single enclosure. Such an enclosure may have a screw-in base thatallows for mechanical attachment to standard Edison sockets and provideelectrical connections to the neutral side of the AC power source andeither the hot side of the AC power source or the dimmed-hot terminal ofthe dimmer switch (e.g., for receipt of the phase-control voltage). Theload regulation circuit is operable to control the intensity of thehigh-efficiency light source to the desired intensity in response to theconduction time of the bidirectional semiconductor switch of the dimmerswitch.

However, the load regulation devices for the high-efficiency lightsources may have high input impedances or input impedances that vary inmagnitude throughout a half-cycle. Therefore, when a prior-art forwardphase-control dimmer switch is coupled between the AC power source andthe load regulation device for the high-efficiency light source, theload control device may not be able to conduct enough current to exceedthe rated latching and/or holding currents of the thyristor. Inaddition, when a prior-art reverse phase-control dimmer switch iscoupled between the AC power source and the load regulation device, themagnitude of the charging current of the power supply may be greatenough to cause the load regulation device to illuminate the controlledhigh-efficiency light source to a level that is perceptible by the humaneye when the light source should be off.

The impedance characteristics of the load regulation device maynegatively affect the magnitude of the phase-control voltage received bythe load regulation device, such that the conduction time of thereceived phase-control voltage is different from the actually conductiontime of the bidirectional semiconductor switch of the dimmer switch(e.g., if the load regulation device has a capacitive impedance).Therefore, the load regulation device may control the intensity of thehigh-efficiency light source to an intensity that is different than thedesired intensity as directed by the dimmer switch. In addition, thecharging current of the power supply of the dimmer switch may build upcharge at the input of a load regulation device having a capacitiveinput impedance, thus negatively affecting the low-end intensity thatmay be achieved.

Therefore, there exists a need for a two-wire load control device thatmay be coupled between an AC power source and a load regulation devicefor a high-efficiency light source and is able to properly control theintensity of the high-efficiency light source.

SUMMARY OF THE INVENTION

According to an embodiment of the present invention, a two-wire loadcontrol device (such as, a dimmer switch) is able to control the amountof power delivered from an AC power source to an electrical load (suchas, a high-efficiency lighting load) and has substantially no minimumload requirement. The dimmer switch includes a bidirectionalsemiconductor switch that is adapted to be coupled in series electricalconnection between the AC power source and the electrical load forconducting a load current from the AC power source to the electricalload. The bidirectional semiconductor switch has a control input forrendering the bidirectional semiconductor switch conductive andnon-conductive. The bidirectional semiconductor switch is operable to berendered conductive each half-cycle and to remain conductive independentof the magnitude of the load current conducted through bidirectionalsemiconductor switch.

The dimmer switch also comprises a control circuit conducts a controlcurrent through the load so as to generate a gate drive signal that isoperatively coupled to the control input of the bidirectionalsemiconductor switch. The control circuit receives a signalrepresentative of a voltage developed across the bidirectionalsemiconductor switch, and is operable to determine a half-cycle starttime near the beginning of a half-cycle of the AC power source inresponse to the signal representative of the voltage developed acrossthe bidirectional semiconductor switch. The control circuit drives thegate drive signal to a first magnitude to render the bidirectionalsemiconductor switch conductive after a first variable amount of timehas elapsed since the half-cycle start time, and maintains the gatedrive signal at the first magnitude after the bidirectionalsemiconductor switch is rendered conductive, such that the bidirectionalsemiconductor switch remains conductive independent of the magnitude ofthe load current conducted through the bidirectional semiconductorswitch. The control circuit drives the gate drive signal to a secondmagnitude to render the bidirectional semiconductor switchnon-conductive after a second fixed amount of time has elapsed since thehalf-cycle start time. The control circuit further controls the secondfixed amount of time to be approximately equal during each half-cycle ofthe AC power source, and varies the first variable amount of time inresponse to the desired amount of power to be delivered to the load tothus control the amount of power delivered to the load to the desiredamount.

According to another embodiment of the present invention, a load controldevice for controlling the amount of power delivered from an AC powersource to an electrical load to a desired amount of power comprises abidirectional semiconductor switch adapted to be coupled in serieselectrical connection between the AC power source and the electricalload for conducting a load current from the AC power source to theelectrical load, and a control circuit including a timing circuit forgenerating a timing signal and a drive circuit for rendering thebidirectional semiconductor switch conductive each half-cycle inresponse to the magnitude of the timing signal, so as to control theamount of power delivered to the electrical load to the desired amount.The control circuit is operable to conduct a control current through theload in order to render the bidirectional semiconductor switchconductive and non-conductive each half-cycle of the AC power source.The timing circuit starts to generate the timing signal at a start timeshortly after a zero-crossing of the AC power source, and the timingsignal increases in magnitude with respect to time. The timing circuitis operable to continue generating the timing signal after thebidirectional semiconductor switch is rendered conductive eachhalf-cycle, such that the drive circuit continues to render thebidirectional semiconductor switch conductive and the bidirectionalsemiconductor switch remains conductive independent of the magnitude ofthe load current conducted through the bidirectional semiconductorswitch.

According to another embodiment of the present invention, the drivecircuit may be operable to render the bidirectional semiconductor switchconductive when the magnitude of the timing signal exceeds a variablethreshold representative of the desired amount of power to be deliveredto the load. In addition, the timing circuit may stop generating thetiming signal after a fixed amount of time has elapsed since the starttime in order to render the bidirectional semiconductor switchnon-conductive.

In addition, the present invention also provides a control circuit for atwo-wire load control device for controlling the amount of powerdelivered from an AC power source to an electrical load. The controlcircuit comprises a timing circuit for generating a timing signal thatincreases in magnitude with respect to time, and a drive circuit forreceiving the timing signal and generating a gate drive signal that isoperatively coupled to a control input of a bidirectional semiconductorswitch of the load control device. The control circuit conducts acontrol current through the load to enable the timing circuit togenerate the timing signal and the drive circuit to generate the gatedrive signal. The timing circuit starts to generate the timing signal ata start time shortly after a zero-crossing of the AC power source, andceases to generate the timing signal after a fixed amount of time haselapsed since the start time. The drive circuit drives the gate drivesignal to a first magnitude to render the bidirectional semiconductorswitch conductive when the magnitude of the timing signal exceeds atrigger threshold, maintains the gate drive signal at the firstmagnitude after the bidirectional semiconductor switch is renderedconductive, and drives the gate drive signal to a second magnitude torender the bidirectional semiconductor switch non-conductive when thetiming circuit ceases generating the timing signal, such that the gatedrive signal is controlled to the first magnitude for a conduction time.The conduction time of the gate drive signal has a length that is notdependent upon the length of the fixed amount of time that the timingcircuit generates the timing signal.

A timing circuit for generating a timing signal in a load control deviceto determine for controlling the amount of power delivered from an ACpower source to an electrical load is also described herein. The timingsignal is used to determine when a bidirectional semiconductor switch ofthe load control device is rendered conductive and non-conductive. Thetiming circuit comprises a constant ramp circuit for generating thetiming signal (which increases in magnitude with respect to time at aconstant rate), a reset circuit coupled to the timing signal forstarting to generate the timing signal at a start time shortly after azero-crossing of the AC power source, and a one-shot circuit coupled tothe timing signal for ceasing to generate the timing signal prior to theend of the present half-cycle after a fixed amount of time has elapsedsince the start time. A dead time exists between the time when theone-shot circuit ceases to generate the timing signal during the presenthalf-cycle and the time when the reset circuit starts to generate thetiming signal at the start time during the next, subsequent half-cycle.

As further described herein, a lighting control system adapted to becoupled to an AC power source comprises a high-efficiency lighting loadincluding a high-efficiency light source and a load regulation device,and a two-wire dimmer switch adapted to be coupled between the AC powersource and the high-efficiency lighting load. The load regulation deviceis electrically coupled to the high-efficiency light source forcontrolling the amount of power delivered to the high-efficiency lightsource, and is characterized by a capacitive impedance. The dimmerswitch comprises a bidirectional semiconductor switch adapted to becoupled in series electrical connection between the AC power source andthe high-efficiency lighting load for conducting a load current from theAC power source to the high-efficiency lighting load. The dimmer switchfurther comprises a control circuit operable to conduct a controlcurrent through the high-efficiency lighting load in order to render thebidirectional semiconductor switch conductive each half-cycle of the ACpower source. The bidirectional semiconductor switch remains conductiveindependent of the magnitude of the load current conducted through thebidirectional semiconductor switch, and is operable to conduct the loadcurrent to and from the high-efficiency lighting load during a singlehalf-cycle of the AC power source. According to another embodiment ofthe present invention, the dimmer switch may have electrical connectionsconsisting of a hot terminal adapted to be coupled to the AC powersource and a dimmed-hot terminal adapted to be coupled to thehigh-efficiency lighting load.

According to yet another embodiment of the present invention, a methodfor controlling the amount of power delivered from an AC power source toan electrical load to a desired amount of power comprises: (1)conducting a load current from the AC power source to the electricalload; (2) controllably rendering a bidirectional semiconductor switchconductive and non-conductive so as control the load current and theamount of power delivered to the load; (3) receiving a signalrepresentative of a voltage developed across the bidirectionalsemiconductor switch; (4) determining a half-cycle start time near thebeginning of a half-cycle of the AC power source in response to thesignal representative of the voltage developed across the bidirectionalsemiconductor switch; (5) conducting a control current through the loadso as to generate a gate drive signal that is operatively coupled to acontrol input of the bidirectional semiconductor switch; (6) driving thegate drive signal to a first magnitude to render the bidirectionalsemiconductor switch conductive after a first variable amount of timehas elapsed since the half-cycle start time; (7) maintaining the gatedrive signal at the first magnitude after the bidirectionalsemiconductor switch is rendered conductive, such that the bidirectionalsemiconductor switch remains conductive independent of the magnitude ofthe load current conducted through the bidirectional semiconductorswitch; (8) driving the gate drive signal to a second magnitude torender the bidirectional semiconductor switch non-conductive after asecond fixed amount of time has elapsed since the half-cycle start time;(9) controlling the second fixed amount of time to be approximatelyequal during each half-cycle of the AC power source; and (10) varyingthe first variable amount of time in response to the desired amount ofpower to be delivered to the load to thus control the amount of powerdelivered to the load to the desired amount.

According to another aspect of the present invention, a two-wire loadcontrol device for controlling the amount of power delivered from an ACpower source to an electrical load to a desired amount of powercomprises a bidirectional semiconductor switch, which is operable to berendered conductive and to remain conductive independent of themagnitude of a load current conducted through semiconductor switch. Thebidirectional semiconductor switch may comprise, for example, twofield-effect transistors coupled in anti-series connection. Thebidirectional semiconductor switch is adapted to be coupled in serieselectrical connection between the AC power source and the electricalload for conducting a load current from the AC power source to theelectrical load. The dimmer switch includes an analog control circuit,such as, for example, a timing circuit, which generates a timing voltagethat increases in magnitude with respect to time. The dimmer switch alsocomprises a drive circuit that receives the timing voltage and rendersthe bidirectional semiconductor switch conductive and non-conductiveeach half-cycle, so as to control the amount of power delivered to theelectrical load to the desired amount.

Other features and advantages of the present invention will becomeapparent from the following description of the invention that refers tothe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will now be described in greater detail in the followingdetailed description with reference to the drawings in which:

FIG. 1 is a simplified block diagram of a lighting control systemincluding a two-wire analog dimmer switch for controlling the intensityof an LED light source according to a first embodiment of the presentinvention;

FIG. 2 is a simplified block diagram of the dimmer switch of FIG. 1according to the first embodiment of the present invention;

FIGS. 3A and 3B show example waveforms illustrating the operation of thedimmer switch of FIG. 1 according to the first embodiment of the presentinvention;

FIG. 4 is a simplified schematic diagram of the dimmer switch of FIG. 2according to the first embodiment of the present invention;

FIG. 5 is a simplified schematic diagram of a timing circuit of thedimmer switch of FIG. 2;

FIG. 6 is a simplified schematic diagram of a dimmer switch according toa second embodiment of the present invention;

FIG. 7 is a simplified block diagram of a reverse-phase control dimmerswitch according to a third embodiment of the present invention;

FIG. 8 is a simplified timing diagram showing examples of waveformsillustrating the operation of the dimmer switch of FIG. 7 according tothe third embodiment of the present invention;

FIG. 9 is a simplified schematic diagram of the dimmer switch of FIG. 7according to the third embodiment of the present invention;

FIG. 10 is a simplified schematic diagram of a dimmer switch accordingto a fourth embodiment of the present invention;

FIG. 11 is a simplified schematic diagram of a dimmer switch accordingto a fifth embodiment of the present invention;

FIG. 12 is a simplified timing diagram showing examples of waveformsillustrating the operation of the dimmer switch of FIG. 11 according tothe fifth embodiment of the present invention;

FIG. 13 is a simplified schematic diagram of a dimmer switch accordingto a sixth embodiment of the present invention;

FIG. 14 is a simplified schematic diagram of a dimmer switch accordingto a seventh embodiment of the present invention;

FIG. 15 is a simplified schematic diagram of a dimmer switch accordingto an eighth embodiment of the present invention;

FIG. 16 is a simplified schematic diagram of a dimmer switch having adigital control circuit according to a ninth embodiment of the presentinvention;

FIG. 17 is a simplified flowchart of a switch procedure executed by amicroprocessor of the dimmer switch of FIG. 16 according to the ninthembodiment of the present invention; and

FIG. 18 is a simplified flowchart of a control procedure periodicallyexecuted by the microprocessor of the dimmer switch of FIG. 16 accordingto the ninth embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The foregoing summary, as well as the following detailed description ofthe preferred embodiments, is better understood when read in conjunctionwith the appended drawings. For the purposes of illustrating theinvention, there is shown in the drawings an embodiment that ispresently preferred, in which like numerals represent similar partsthroughout the several views of the drawings, it being understood,however, that the invention is not limited to the specific methods andinstrumentalities disclosed.

FIG. 1 is a simplified block diagram of a lighting control system 10including a “two-wire” dimmer switch 100 for controlling the amount ofpower delivered to a high-efficiency lighting load 101 including a loadregulation device, e.g., a light-emitting diode (LED) driver 102, and ahigh-efficiency light source, e.g., an LED light source 104 (or “lightengine”). The dimmer switch 100 has a hot terminal H coupled to analternating-current (AC) power source 105 for receiving an AC mains linevoltage V_(AC), and a dimmed-hot terminal DH coupled to the LED driver102. The dimmer switch 100 does not require a direct connection to theneutral side N of the AC power source 105. The dimmer switch 100generates a phase-control voltage V_(PC) (e.g., a dimmed-hot voltage) atthe dimmed-hot terminal DH and conducts a load current I_(LOAD) throughthe LED driver 102. The dimmer switch 100 may either use forwardphase-control dimming or reverse phase-control dimming techniques togenerate the phase-control voltage V_(PC).

As defined herein, a “two-wire” dimmer switch or load control devicedoes not require a require a direct connection to the neutral side N ofthe AC power source 105. In other words, all currents conducted by thetwo-wire dimmer switch must also be conducted through the load. Atwo-wire dimmer switch may have only two terminals (i.e., the hotterminal H and the dimmed hot terminal DH as shown in FIG. 1).Alternatively, a two-wire dimmer switch (as defined herein) couldcomprise a three-way dimmer switch that may be used in a three-waylighting system and has at least three load terminals, but does notrequire a neutral connection. In addition, a two-wire dimmer switch maycomprise an additional connection that provides for communication with aremote control device (for remotely controlling the dimmer switch), butdoes not require the dimmer switch to be directly connected to neutral.

The LED driver 102 and the LED light source 104 may be both includedtogether in a single enclosure, for example, having a screw-in baseadapted to be coupled to a standard Edison socket. When the LED driver102 is included with the LED light source 104 in the single enclosure,the LED driver only has two electrical connections: to the dimmer switch100 for receiving the phase-control voltage V_(PC) and to the neutralside N of the AC power source 105. The LED driver 102 comprises arectifier bridge circuit 106 that receives the phase-control voltageV_(PC) and generates a bus voltage V_(BUS) across a bus capacitorC_(BUS). The LED driver 102 further comprises a load control circuit 107that receives the bus voltage V_(BUS) and controls the intensity of theLED light source 104 in response to the phase-control signal V_(PC).Specifically, the load control circuit 107 of the LED driver 102 isoperable to turn the LED light source 104 on and off and to adjust theintensity of the LED light source to a target intensity L_(TRGT) (i.e.,a desired intensity) in response to the phase-control signal V_(PC). Thetarget intensity L_(TRGT) may range between a low-end intensity L_(LE)(e.g., approximately 1%) and a high-end intensity L_(HE) (e.g.,approximately 100%). The LED driver 102 may also comprise a filternetwork 108 for preventing noise generated by the load control circuit107 from being conducted on the AC mains wiring. An example of the LEDdriver 102 is described in greater detail in U.S. patent applicationSer. No. 12/813,908, filed Jun. 11, 2009, entitled LOAD CONTROL DEVICEFOR A LIGHT-EMITTING DIODE LIGHT SOURCE, the entire disclosure of whichis hereby incorporated by reference.

In addition, the LED driver 102 may comprise an artificial load circuit109 for conducting current (in addition to the load current I_(LOAD))through the dimmer switch 100. Accordingly, if the dimmer switch 100includes a triac for generating the phase-control voltage V_(PC), theartificial load circuit 109 may conduct enough current to ensure thatthe magnitude of the total current conducted through the triac of thedimmer switch 100 exceeds the rated latching and holding currents of thetriac. In addition, the artificial load circuit 109 may conduct a timingcurrent if the dimmer switch 100 comprises a timing circuit and mayconduct a charging current if the dimmer switch comprises a powersupply, such that these currents need not be conducted through the loadcontrol circuit 107 and do not affect the intensity of the LED lightsource 104.

The artificial load circuit 109 may simply comprise a constant impedancecircuit (e.g., a resistor) or may comprise a current source circuit.Alternatively, the artificial load circuit 109 may be controllable, suchthat the artificial load circuit may be enabled and disabled to thusselectively conduct current through the dimmer switch 100. In addition,the artificial load circuit 109 may be controlled to conduct differentamounts of current depending upon the magnitude of the AC mains linevoltage V_(AC), the present time during a half-cycle of the AC mainsline voltage, or the present operating mode of the LED driver 102.Examples of artificial load circuits are described in greater detail incommonly-assigned U.S. patent application Ser. No. 12/438,587, filedAug. 5, 2009, entitled VARIABLE LOAD CIRCUITS FOR USE WITH LIGHTINGCONTROL DEVICES, and U.S. patent application Ser. No. 12/950,079, filedNov. 19, 2010, entitled CONTROLLABLE-LOAD CIRCUIT FOR USE WITH A LOADCONTROL DEVICE, the entire disclosures of which are hereby incorporatedby reference.

Alternatively, the high-efficiency light source could comprise a compactfluorescent lamp (CFL) and the load regulation device could comprise anelectronic dimming ballast. In addition, the dimmer switch 100 couldalternatively control the amount of power delivered to other types ofelectrical loads, for example, by directly controlling a lighting loador a motor load. An example of a screw-in light source having afluorescent lamp and an electronic dimming ballast is described ingreater detail in U.S. patent application Ser. No. 12/704,781, filedFeb. 12, 2010, entitled HYBRID LIGHT SOURCE, the entire disclosure ofwhich is hereby incorporated by reference.

The dimmer switch 100 comprises a user interface having a rocker switch116 and an intensity adjustment actuator 118 (e.g., a slider knob asshown in FIG. 1). The rocker switch 116 allows for turning on and offthe LED light source 104, while the intensity adjustment actuator 118allows for adjustment of the target intensity L_(TRGT) of the LED lightsource 104 from the low-end intensity L_(LE) to the high-end intensityL_(HE). Examples of user interfaces of dimmer switches are described ingreater detail in commonly-assigned U.S. patent application Ser. No.12/363,258, filed Jan. 30, 2009, entitled LOAD CONTROL DEVICE HAVING AVISUAL INDICATION OF ENERGY SAVINGS AND USAGE INFORMATION, the entiredisclosure of which is hereby incorporated by reference.

FIG. 2 is a simplified block diagram of the dimmer switch 100 accordingto a first embodiment of the present invention. FIGS. 3A and 3B showexample waveforms illustrating the operation of the dimmer switch 100according to the first embodiment of the present invention. The dimmerswitch 100 comprises a bidirectional semiconductor switch 110 coupledbetween the hot terminal H and the dimmed hot terminal DH for generatingthe phase-control voltage V_(PC) (as shown in FIGS. 3A and 3B) andcontrolling of the amount of power delivered to the LED driver 102. Thebidirectional semiconductor switch 110 comprises a control input (e.g.,a gate), which may receive control signals for rendering thebidirectional semiconductor switch conductive and non-conductive. Thebidirectional semiconductor switch 110 may comprise a single device,such as a triac, or a combination of devices, such as, two field-effecttransistors (FETs) coupled in anti-series connection. According to thefirst embodiment of the present invention, the phase-control voltageV_(PC) comprises a forward phase-control voltage. In other words, thephase-control voltage V_(PC) has a magnitude of approximately zero voltsat the beginning of each half-cycle during a non-conduction time T_(NC),and has a magnitude equal to approximately the magnitude of the AC linevoltage V_(AC) of the AC power source 105 during the rest of thehalf-cycle, i.e., during a conduction time T_(CON).

The dimmer switch 100 comprises a mechanical air-gap switch S112electrically coupled to the hot terminal H and in series with thebidirectional semiconductor switch 110, such that the LED light source104 is turned off when the switch is open. When the air-gap switch S112is closed, the dimmer switch 100 is operable to control thebidirectional semiconductor switch 110 to control the amount of powerdelivered to the LED driver 102. The air-gap switch S112 is mechanicallycoupled to the rocker switch 116 of the user interface of the dimmerswitch 100, such that the switch may be opened and closed in response toactuations of the rocker switch. The dimmer switch 100 further comprisesa rectifier circuit 114 coupled across the bidirectional semiconductorswitch 110 and operable to generate a rectified voltage V_(RECT) (i.e.,a signal representative of the voltage developed across thebidirectional semiconductor switch).

According to the first embodiment, the dimmer switch 100 comprises ananalog control circuit 115 including a power supply 120, a constant-rateone-shot timing circuit 130, and a variable-threshold trigger circuit140 (i.e., a gate drive circuit). The control circuit 115 receives therectified voltage V_(RECT) from the rectifier circuit 114 and conducts acontrol current I_(CNTL) through the load (i.e., the LED driver 102) inorder to generate a drive voltage V_(DR) for controlling thebidirectional semiconductor switch 110 to thus adjust the intensity ofthe LED light source 104 in response to the intensity adjustmentactuator 118. The power supply 120 of the control circuit 115 conducts acharging current I_(CHRG) through the LED driver 102 in order togenerate a supply voltage V_(CC) (e.g., approximately 11.4 volts). Thecharging current I_(CHRG) of the power supply makes up a portion of thecontrol current I_(CNTL) of the control circuit 115.

The timing circuit 130 receives the supply voltage V_(CC) and generatesa timing voltage V_(TIM) (i.e., a timing signal), which comprises a rampsignal having a constant rate of increasing magnitude (i.e., a constantpositive slope) as shown in FIGS. 3A and 3B. When the bidirectionalsemiconductor switch 110 is non-conductive at the beginning of eachhalf-cycle, the timing circuit 130 also receives the rectified voltageV_(RECT) and is able to derive zero-crossing timing information from thevoltage developed across the LED driver 102 (i.e., from the controlcurrent I_(CNTL) conducted through the LED driver 102). The timingvoltage V_(TIM) begins increasing from approximately zero volts shortlyafter the zero-crossings of the AC line voltage V_(AC) (i.e., shortlyafter the beginning of each half-cycle as shown at times t₁, t₄ in FIGS.3A and 3B) and continues increasing at the constant rate. After a fixedamount of time T_(TIM) has elapsed since the timing voltage V_(TIM)started increasing from zero volts during the present half-cycle, thetiming voltage V_(TIM) is driven to approximately zero volts near thenext zero-crossing (i.e., near the end of the present half-cycle asshown at time t₃ in FIGS. 3A and 3B). Since the timing voltage V_(TIM)increases in magnitude at the constant rate for the fixed amount of timeT_(TIM) each half-cycle, the timing voltage V_(TIM) is essentiallyidentical during each half-cycle as shown in FIGS. 3A and 3B.

Referring back to FIG. 2, the variable-threshold trigger circuit 140receives the timing voltage V_(TIM) from the timing circuit 130, andgenerates a drive voltage V_(DR) (i.e., a gate drive voltage) forcontrolling the bidirectional semiconductor switch 110 to thus adjustthe intensity of the LED light source 104 in response to actuations ofthe intensity adjustment actuator 118. The trigger circuit 140 ischaracterized by a variable threshold (i.e., a variable thresholdvoltage V_(TH) shown in FIGS. 3A and 3B) that may be adjusted inresponse to the intensity adjustment actuator 118 of the user interfaceof the dimmer switch 100.

A gate coupling circuit 150 couples the drive voltage V_(DR) to the gateof the bidirectional semiconductor switch 110 for thus rendering thebidirectional semiconductor switch 110 conductive and non-conductive inresponse to the magnitude of the variable threshold voltage V_(TH). Whenthe magnitude of the timing voltage V_(TIM) exceeds the magnitude of avariable threshold voltage V_(TH) each half-cycle (as shown at firingtimes t₂, t₅ in FIGS. 3A and 3B), the trigger circuit 140 is operable todrive the drive signal V_(DR) to a first magnitude (e.g., approximatelyzero volts as shown in FIGS. 3A and 3B) to thus render the bidirectionalsemiconductor switch 110 conductive each half-cycle (as will bedescribed in greater detail below with reference to FIG. 4). The drivesignal V_(DR) is then driven to a second magnitude (e.g., approximatelythe supply voltage V_(CC) as shown in FIGS. 3A and 3B) to render thebidirectional semiconductor switch 110 non-conductive when the timingvoltage V_(TIM) is controlled to approximately zero volts shortly beforethe next zero-crossing. The variable threshold voltage V_(TH) is shownat two different magnitudes in FIGS. 3A and 3B, which results in thedrive signal V_(DR) being driven low to zero volts (and thus renderingthe bidirectional semiconductor switch 110 conductive) for differentamounts of time.

As shown in FIGS. 3A and 3B, the control circuit 115 of the dimmerswitch 100 is operable to provide a constant gate drive to thebidirectional semiconductor switch 110 by maintaining the drive voltageV_(DR) low for the remainder of the half-cycle after the bidirectionalsemiconductor switch 110 is rendered conductive (as shown at firingtimes t₂, t₅). Accordingly, the bidirectional semiconductor switch 110will remain conductive independent of the magnitude of the load currentI_(LOAD) conducted through the bidirectional semiconductor switch andthe LED driver 102. When the bidirectional semiconductor switch 110 isconductive and the magnitude of the phase control voltage V_(PC) isgreater than approximately the magnitude of the bus voltage V_(BUS) ofthe LED driver 102, the LED driver 102 will begin to conduct the loadcurrent I_(LOAD) through the bidirectional semiconductor switch. Sincethe bus capacitor C_(BUS) of the LED driver 102 may charge quickly, themagnitude of the load current I_(LOAD) may quickly peak before subsidingdown to a substantially small magnitude (e.g., approximately zero amps).As previously mentioned, the bidirectional semiconductor switch 110 willremain conductive independent of the magnitude of the load currentI_(LOAD) because the control circuit 115 is providing constant gatedrive to the bidirectional semiconductor switch. In addition to quicklyincreasing and decreasing in magnitude, the load current I_(LOAD) mayalso change direction after the bidirectional semiconductor switch 110is rendered conductive. Therefore, the bidirectional semiconductorswitch 110 is also operable to conduct current in both directions (i.e.,to and from the LED driver 102) after the bidirectional semiconductorswitch is rendered conductive during a single half-cycle, therebyallowing any capacitors in the filter network 108 of the LED driver 102to follow the magnitude of the AC line voltage V_(AC) of the AC powersource 105.

FIG. 4 is a simplified schematic diagram of the dimmer switch 100. Asshown in FIG. 4, the bidirectional semiconductor switch 110 of thedimmer switch 100 of the first embodiment is implemented as a triac110′, but may alternatively be implemented as one or moresilicon-controlled rectifiers (SCRs), or any suitable thyristor. Whilenot shown in FIG. 4, a choke inductor may be coupled in series with thetriac 110′, and a filter circuit (such as a filter capacitor) may becoupled between the hot terminal H and the dimmed hot terminal DH (i.e.,in parallel with the triac) to prevent noise generated by the switchingof the triac from being conducted on the AC mains wiring. The rectifiercircuit 114 comprises a full-wave rectifier bridge having four diodesD114A, D114B, D114C, D114D. The rectifier bridge of the rectifiercircuit 114 has AC terminals coupled in series between the hot terminalH and the dimmed hot terminal DH, and DC terminals for providing therectified voltage V_(RECT) to the timing circuit 130 when the triac 110′is non-conductive and a voltage is developed across the dimmer switch100. The control circuit 115 conducts the control current I_(CNTL)through the rectifier circuit 114 and the LED driver 102. Accordingly,the total current conducted through the LED driver 102 each half-cycleis the sum of the load current I_(LOAD) conducted through thebidirectional semiconductor switch 110, the control current I_(CNTL)conducted through the control circuit 115 of the dimmer switch 100, andany leakage current conducted through the filter circuit (that may becoupled between the hot terminal H and the dimmed hot terminal DH).

As shown in FIG. 4, the power supply 120 comprises, for example, apass-transistor circuit that generates the supply voltage V_(CC). Thepass-transistor circuit comprises an NPN bipolar junction transistorQ122 having a collector coupled to receive the rectifier voltageV_(RECT) through a resistor R124 (e.g., having a resistance ofapproximately 100 kΩ). The base of the transistor Q122 is coupled to therectifier voltage V_(RECT) through a resistor R125 (e.g., having aresistance of approximately 150 kΩ), and to circuit common through azener diode Z126 (e.g., having a break-over voltage of approximately 12volts). The power supply 120 further comprises a storage capacitor C128,which is able to charge through the transistor Q122 to a voltage equalto approximately the break-over voltage of the zener diode Z126 minusthe base-emitter drop of the transistor Q122. The storage capacitor C128has, for example, a capacitance of approximately 10 μF, and operates tomaintain the supply voltage V_(CC) at an appropriate magnitude (i.e.,approximately 11.4 volts) to allow the timing circuit 120 to generatethe timing voltage V_(TIM) and the gate coupling circuit 150 to continuerendering the triac 110′ conductive after the firing times eachhalf-cycle.

The timing circuit 130 comprises a constant ramp circuit 160, a one-shotlatch circuit 170, and a reset circuit 180. The constant ramp circuit160 receives the supply voltage V_(CC) and causes the timing voltageV_(TIM) to increase in magnitude at the constant rate. The reset circuit180 receives the rectified voltage V_(RECT) and is coupled to the timingvoltage V_(TIM), such that the reset circuit is operable to start thetiming voltage V_(TIM) increasing in magnitude from approximately zerovolts shortly after the beginning of each half-cycle at a half-cyclestart time (e.g., times t₁, t₄ in FIGS. 3A and 3B). Specifically, thereset circuit 180 is operable to enable the timing voltage V_(TIM)(i.e., to start the increase of the magnitude of the timing voltageV_(TIM)) in response to a positive-going transition of the rectifiedvoltage V_(RECT) across a reset threshold V_(RST) that remains above thereset threshold V_(RST) for at least a predetermined amount of time. Theone-shot latch circuit 170 provides a latch voltage V_(LATCH) to thereset circuit 180 to prevent the reset circuit 180 from resetting thetiming voltage V_(TIM) until the end of the half-cycle, thus ensuringthat the reset circuit only restarts the generation of the timingvoltage once each half-cycle.

The one-shot latch circuit 170 stops the generation of the timingvoltage V_(TIM) by controlling the magnitude of the timing voltageV_(TIM) to approximately 0.6 volts at the end of the fixed amount oftime from when the reset circuit 180 enabled the timing voltage V_(TIM)(e.g., near the end of the half-cycle at time t₃ in FIGS. 3A and 3B).After the one-shot latch circuit 170 controls the magnitude of thetiming voltage V_(TIM) to approximately 0.6 volts, the reset circuit 180is once again able to enable the generation of the timing voltageV_(TIM) after the beginning of the next half-cycle (i.e., at time t₄ inFIGS. 3A and 3B). As a result, a dead time T_(DT) exists between thetime when the one-shot latch circuit 170 drives the timing voltageV_(TIM) to approximately 0.6 volts and the reset circuit 180 enables thegeneration of the timing voltage V_(TIM) by controlling the magnitude ofthe timing voltage V_(TIM) down to approximately zero volts.

The variable-threshold trigger circuit 140 comprises a comparator U142having an inverting input that receives the timing voltage V_(TIM) fromthe timing circuit 130. The variable-threshold trigger circuit 140 alsocomprises a potentiometer R144 that is mechanically coupled to theslider knob of the intensity adjustment actuator 118. The potentiometerR144 has a resistive element coupled between the supply voltage V_(CC)and circuit common and a wiper terminal that generates the variablethreshold voltage V_(TH). The variable threshold voltage V_(TH)comprises a DC voltage that varies in magnitude in response to theposition of the slider knob of the intensity adjustment actuator 118 andis provided to a non-inverting input of the comparator U142. The drivevoltage V_(DR) is generated at an output of the comparator U142 and isprovided to the gate coupling circuit 150 for rendering the triac 110′conductive and non-conductive. The gate coupling circuit 150 comprisesan opto-coupler U152 having an input photodiode, which is coupledbetween the supply voltage V_(CC) and the output of the comparator U142and in series with a resistor R154 (e.g., having a resistance ofapproximately 8.2 kΩ). The opto-coupler U152 has an output phototriacthat is coupled between the hot terminal H and the gate of the triac110′ and in series with a resistor R156 (e.g., having a resistance ofapproximately 100Ω).

When the magnitude of the timing voltage V_(TIM) is below the magnitudeof the variable threshold voltage V_(TH), the magnitude of the drivevoltage V_(DR) at the output of the comparator U142 of thevariable-threshold trigger circuit 140 remains high at approximately thesupply voltage V_(CC), such that the triac 110′ remains non-conductive.When the magnitude of the timing voltage V_(TIM) increases above thevariable threshold voltage V_(TH), the comparator U142 drives the drivevoltage V_(DR) low to approximately circuit common, such that the inputphotodiode of the opto-coupler U152 conducts a drive current I_(DR). Asa result, the output phototriac of the opto-coupler U152 is renderedconductive, thus also rendering the triac 110′ conductive. Accordingly,the drive voltage V_(DR) is driven low to render the triac 110′conductive after a variable amount of time has elapsed since thehalf-cycle start time (i.e., the non-conduction time T_(NC) as shown inFIGS. 3A and 3B), where the variable amount of time is adjusted inresponse to intensity adjustment actuator 118 and the variable thresholdvoltage V_(TH). After the triac 110′ is rendered conductive eachhalf-cycle, the timing circuit 130 continues to generate the timingvoltage V_(TIM). Thus, the magnitude of the timing voltage V_(TIM)remains above the variable threshold voltage V_(TH) and the triac 110′remains conductive until approximately the end of the half-cycle whenthe one-shot latch circuit 170 drives the timing voltage toapproximately zero volts.

According to the first embodiment of the present invention, the latchcircuit 170 is operable to control the timing voltage V_(TIM) toapproximately zero volts (thus controlling the magnitude of the drivevoltage V_(DR) high to approximately the supply voltage V_(CC)) shortlybefore the end of the present half-cycle (as shown at time t₃ in FIGS.3A and 3B). Accordingly, the length of the timing voltage V_(TIM) (i.e.,the fixed amount of time T_(TIM)) is slightly smaller than the lengthT_(HC) of each half-cycle. The dead time T_(DT) (or “blanking pulse”) inthe timing voltage V_(TIM) at the end of the half-cycle allows the triac110′ to commutate off (i.e., become non-conductive) when the magnitudeof the load current I_(LOAD) through the triac reduces to approximatelyzero amps at the end of the half-cycle.

Because the LED driver 102 may have a capacitive input impedance, themagnitude of the phase-control voltage V_(PC) may not quickly decreaseto zero volts near the zero-crossing of the AC mains lines voltage VACafter the triac 110′ becomes non-conductive at the end of eachhalf-cycle. Therefore, according to the first embodiment of the presentinvention, the reset circuit 180 only starts the timing voltage V_(TIM)after a zero-crossing of the AC mains lines voltage V_(AC), i.e., inresponse to the magnitude of the rectified voltage V_(RECT) exceedingthe reset threshold V_(RST) when the rectified voltage is increasing inmagnitude. The reset circuit 180 is prevented from resetting the timingvoltage V_(TIM) in response to the magnitude of the rectified voltageV_(RECT) dropping below the reset threshold V_(RST), which may or maynot happen each half-cycle due to the capacitive input impedance of theLED driver 102.

FIG. 5 is a simplified schematic diagram of the timing circuit 130. Theconstant ramp circuit 160 receives the supply voltage V_(CC) andgenerates the timing voltage V_(TIM) across a timing capacitor C162(e.g., having a capacitance of approximately 50 nF). The constant rampcircuit 160 comprises a constant current source for conducting aconstant timing current I_(TIM) through the timing capacitor C162, suchthat the timing voltage V_(TIM) has a constant slope. The constantcurrent source circuit comprises a PNP bipolar junction transistor Q164having an emitter coupled to the supply voltage V_(CC) via a resistorR165 (e.g. having a resistance of approximately 10 kΩ). Two diodes D166,D168 are coupled in series between the supply voltage V_(CC) and thebase of the transistor Q164. A resistor R169 is coupled between the baseof the transistor Q164 and circuit common and has, for example, aresistance of approximately 51 kΩ. A voltage having a magnitude ofapproximately the forward voltage drop of the diode D166 (e.g.,approximately 0.6 V) is produced across the resistor R165, such that theresistor conducts the constant timing current I_(TIM) (e.g.,approximately 70 μA) into the capacitor C162. The rate at which themagnitude of the timing voltage V_(TIM) increases with respect to time(i.e., dV_(TIM/dt)) is a function of the magnitude of the timing currentI_(TIM) and the capacitance C_(C162) of the capacitor C162 (i.e.,dV_(TIM)/dt=I_(TIM)/C₁₆₂), and may be equal to, for example,approximately 1.4 V/msec.

The one-shot latch circuit 170 comprises a comparator U172 having aninverting input coupled to the timing voltage V_(TIM). The timingvoltage V_(TIM) is further coupled to an output of the comparator U172via a diode D174. The one-shot latch circuit 170 includes a resistivedivider, which is coupled in series electrical connection between thesupply voltage V_(CC) and circuit common, and comprises two resistorsR175, R176 having, for example, resistances of approximately 100 kΩ and1 MΩ, respectively. The junction of the two resistors R175, R176produces a latch threshold voltage V_(TH-L), which is provided to anon-inverting input of the comparator U172. The non-inverting input ofthe comparator U172 is also coupled to the output via a resistor R178(e.g., having a resistance of approximately 1 kΩ). The latch voltageV_(LATCH) is generated at the output of the comparator U172 and isprovided to the reset circuit 180 as will be described in greater detailbelow.

The reset circuit 180 comprises a first comparator U181 having anon-inverting input that receives the rectified voltage V_(RECT) via theseries combination of a zener diode Z182 and a resistor R183 (e.g.,having a resistance of approximately 100 kΩ). The parallel combinationof a capacitor C184 (e.g., having a capacitance of approximately 1000pF) and a resistor R185 (e.g., having a resistance of approximately 20kΩ) is coupled between the non-inverting input of the comparator U181and circuit common. A zener diode Z186 (e.g., having a break-overvoltage of approximately 12 volts) clamps the magnitude of the voltageproduced between the non-inverting input of the comparator U181 andcircuit common. The reset circuit 180 further comprises a resistivedivider that has two resistors R187, R188 (e.g., having resistances ofapproximately 150 kΩ and 100 kΩ, respectively), and is coupled in serieselectrical connection between the supply voltage V_(CC) and circuitcommon. The junction of the two resistors R187, R188 produces a resetthreshold voltage V_(RST) (e.g., approximately 4.8 V), which is providedto an inverting input of the comparator U181. An output of thecomparator U181 is coupled to the supply voltage V_(CC) via a resistorR189 (e.g., having a resistance of approximately 10 kΩ).

The reset circuit 180 also comprises a second comparator U191 having anon-inverting input coupled to the threshold voltage V_(RST) and anoutput coupled to the timing voltage V_(TIM). The output of thecomparator U181 is coupled to an inverting input of the secondcomparator U191 via a capacitor C190 (e.g., having a capacitance ofapproximately 1000 pF). A resistor R192 (e.g., having a resistance ofapproximately 68 kΩ) and a diode D193 are coupled between the invertinginput of the comparator U191 and circuit common. A FET Q194 is alsocoupled between the inverting input and circuit common. The gate of theFET Q194 is pulled up towards the supply voltage V_(CC) through aresistor R195 (e.g., having a resistance of approximately 100 kΩ), andis coupled to the latch voltage V_(LATCH), such that the FET may berendered conductive and non-conductive in response to the one-shot latchcircuit 170.

When the timing voltage V_(TIM) starts out at approximately zero volts,the inverting input of the comparator U172 of the latch circuit 170 isless than the latch threshold voltage V_(TH-L) (e.g., approximately 10.5V) at the non-inverting input and the output is pulled up towards thesupply voltage V_(CC) via the resistor R195 and the diode D196 of thereset circuit 180. The magnitude of the timing voltage V_(TIM) continuesto increase at the constant rate until the magnitude of timing voltageexceeds the latch threshold voltage V_(TH-L), at which time, thecomparator U172 of the latch circuit 170 drives the output low toapproximately zero volts. At this time, the magnitude of the timingvoltage V_(TIM) is reduced to approximately the forward voltage drop ofthe diode D174 (e.g., approximately 0.6 V). Accordingly, the fixedamount of time T_(TIM) that the timing voltage V_(TIM) is generated eachhalf-cycle is a function of the constant rate at which the magnitude ofthe timing voltage V_(TIM) increases with respect to time dV_(TIM)/dt(i.e., approximately 1.4 V/msec) and the magnitude of the latchthreshold voltage V_(TH-L) (i.e., approximately 10.5 V), such that thefixed amount of time T_(TIM) is approximately 7.5 msec each half-cycle.After the magnitude of the timing voltage V_(TIM) has exceeded the latchthreshold voltage V_(TH-L), the latch threshold voltage V_(TH-L) isreduced to approximately 0.1 V, such that the comparator U172 continuesto drive the output low and the magnitude of the timing voltage V_(TIM)is maintained at approximately 0.6 V.

At the beginning of a half-cycle, the magnitude of the rectified voltageV_(RECT) is below a break-over voltage of the zener diode Z182 of thereset circuit 180 (e.g., approximately 30 V) and the voltage at thenon-inverting input of the first comparator U181 is approximately zerovolts, such that the output of the first comparator is driven lowtowards circuit common. When the magnitude of the rectified voltageV_(RECT) exceeds approximately the break-over voltage of the zener diodeZ182, the capacitor C184 begins to charge until the magnitude of thevoltage at the non-inverting input of the first comparator U181 exceedsthe reset threshold voltage V_(RST). The output of the first comparatorU181 is then driven high towards the supply voltage V_(CC) and thecapacitor C190 conducts a pulse of current into the resistor R192, suchthat the magnitude of the voltage at the inverting input of the secondcomparator U191 exceeds the reset threshold voltage V_(RST), and thesecond comparator pulls the timing voltage V_(TIM) down towards circuitcommon (i.e., the magnitude of the timing voltage is controlled fromapproximately 0.6 volts to zero volts). The magnitude of the voltage atthe inverting input of the comparator U172 of the latch circuit 170 isnow less than the latch threshold voltage V_(TH-L) (i.e., approximately0.1 V), and the comparator stops pulling the timing voltage V_(TIM) downtowards circuit common. In addition, the reset circuit 180 only drivesthe timing voltage V_(TIM) low for a brief period of time (e.g.,approximately 68 μsec) before the capacitor C190 fully charges and thenstops conducting the pulse of current into the resistor R192.Accordingly, the second comparator U191 then stops pulling the timingvoltage V_(TIM) down towards circuit common, thus allowing the timingvoltage to once again begin increasing in magnitude with respect to timeat the constant rate.

After the reset circuit 180 resets the generation of the timing voltageV_(TIM) after the beginning of each half-cycle, the comparator U172 ofthe latch circuit 170 stops pulling the timing voltage V_(TIM) downtowards circuit common and the magnitude of the latch voltage V_(LATCH)is pulled high towards the supply voltage V_(CC) via the resistor R195and the diode D196. At this time, the FET Q194 is rendered conductive,thus maintaining the inverting input of the second comparator U191 lessthan the reset threshold voltage V_(RST). The FET Q194 is renderednon-conductive when the comparator U172 of the one-shot latch circuit170 pulls the timing voltage V_(TIM) low near the end of the half-cycle.Thus, the FET Q194 is rendered conductive for most of each half-cycleand prevents the reset circuit 180 from resetting the generation of thetiming voltage V_(TIM) until after the latch circuit 170 ceases thegeneration of the timing voltage, thereby greatly improving the noiseimmunity of the dimmer switch 100 with respect to impulse noise on theAC line voltage V_(AC).

When the magnitude of the voltage at the non-inverting input of thefirst comparator U181 of the reset circuit 170 exceeds the resetthreshold voltage V_(RST), the output is then driven high towards thesupply voltage V_(CC) and the capacitor C190 charges. The FET Q194 isthen rendered conductive, and the capacitor C190 remains charged. Whenthe magnitude of the rectified voltage V_(RECT) drops below thebreak-over voltage of the zener diode Z182 at the end of each half-cycleand the magnitude of the voltage at the non-inverting input of the firstcomparator U181 drops below the reset threshold voltage V_(RST), thecapacitor C190 discharges through the diode D193 and the output of thefirst comparator U181. However, the magnitude of the voltage at theinverting input of the second comparator U191 remains less than thereset threshold voltage V_(RST), and thus the reset circuit 180 does notreset the generation of the timing voltage V_(TIM) until the magnitudeof the voltage at the non-inverting input of the first comparator U181of the reset circuit 170 rises above the reset threshold voltage V_(RST)at the beginning of the next half-cycle.

Accordingly, the control circuit 115 of the dimmer switch 100 of thefirst embodiment of the present invention conducts a control currentthrough the LED driver 102 and provides constant gate drive to thebidirectional semiconductor switch 110 after the bidirectionalsemiconductor switch is rendered conductive. The control circuit 115 isoperable to derive zero-crossing timing information from the voltagedeveloped across the LED driver 102, and thus from the control currentI_(CNTL) conducted through the LED driver 102. The average magnitude ofthe control current I_(CNTL) conducted through the LED driver 102 isapproximately equal to the sum of the average magnitude of the timingcurrent I_(TIM) and the drive current I_(DR), as well as the othercurrents drawn by the timing circuit 130 and the trigger circuit 140.The control circuit 115 is operable to render the bidirectionalsemiconductor switch 110 conductive each half-cycle in response to thevariable threshold that is representative of the desired intensity ofthe LED light source 104 and to maintain the bidirectional semiconductorswitch conductive until approximately the end of the present half-cycle.As a result, the conduction time T_(CON) of the drive voltage V_(DR)generated by the trigger circuit 140 has a length that is not dependentupon the length of the fixed amount of time I_(TIM) that the timingcircuit 130 generates the timing signal V_(TIM).

FIG. 6 is a simplified schematic diagram of a dimmer switch 200according to a second embodiment of the present invention. Thebidirectional semiconductor switch of the dimmer switch 200 of thesecond embodiment is implemented as two individual switchingtransistors, e.g., FETs Q210A, Q210B, coupled in anti-series connectionbetween the hot terminal H and the dimmed hot terminal DH for control ofthe amount of power delivered to the LED driver 102. The sources of theFETs Q210A, Q210B are coupled together at circuit common. The FETsQ210A, Q210B may comprise metal-oxide semiconductor FETs (MOSFETs) ormay alternatively be replaced by any suitable semiconductor switch, suchas, for example, insulated gate bipolar junction transistors (IGBT). TheFETs Q210A, Q210B have control inputs (i.e., gates) that are coupled toa gate coupling circuit 250, that comprises respective gate resistors8252, 8254 (e.g., each having a resistance of approximately 47Ω) forcoupling a drive voltage V_(DR-INV) to the gates of the FETs. The drivevoltage V_(DR-INV) is the inverse of the drive voltage V_(DR) of thefirst embodiment as shown in FIGS. 3A and 3B. The FETs Q210A, Q210B aresimultaneously controlled to be conductive and non-conductive using theforward phase-control technique, and are operable to be renderedconductive and to remain conductive independent of the magnitude of theload current I_(LOAD) conducted through the FETs.

The dimmer switch 200 comprises a full-wave rectifier bridge thatincludes the body diodes of the two FETs Q210A, Q210B in addition to twodiodes D214A, D214B. The timing circuit 130 of the dimmer switch 200 ofthe second embodiment operates in the same manner as in the firstembodiment. The dimmer switch 200 comprises a variable-threshold triggercircuit 240 that is similar to the variable-threshold trigger circuit140 of the first embodiment. However, the trigger circuit 240 of thesecond embodiment comprises a comparator U242 having a non-invertinginput that receives the timing voltage V_(TIM) and an inverting inputthat receives a variable threshold voltage V_(TH) from a potentiometer8244. The trigger circuit 240 operates to drive the drive voltageV_(DR-INV) high towards the supply voltage V_(CC) to render the FETsQ210A, Q210B conductive, and low towards circuit common to render theFETs non-conductive.

FIG. 7 is a simplified block diagram of a reverse-phase control dimmerswitch 300 according to a third embodiment of the present invention. Asshown in FIG. 7, the bidirectional semiconductor switch is implementedas two FETs Q210A, Q210B coupled in anti-series connection (as in thesecond embodiment). The dimmer switch 100 comprises an analog controlcircuit including a voltage reference circuit 320, a timing circuit 330,and a gate drive circuit 340. The voltage reference circuit 320 includesa pass-transistor circuit 360 and a snap-on circuit 370, and operates togenerate a reference voltage V_(REF) (e.g., approximately 14.4 volts)from the rectified voltage V_(RECT). The timing circuit 330 receives thereference voltage V_(REF) and generates a timing voltage V_(TIM), whichis representative of the target intensity L_(TRGT) of the LED lightsource 104. The gate drive circuit 340 generates a gate voltage V_(G),which is coupled to the gates of the FETs Q210A, Q210B via the gatecoupling circuit 250 for simultaneously rendering the FETs conductiveand non-conductive. According to the third embodiment of the presentinvention, the phase-control voltage V_(PC) generated by the dimmerswitch 300 comprises a reverse phase-control voltage. Accordingly, thegate drive circuit 340 operates to render the FETs Q210A, Q210Bconductive at the beginning of each half-cycle, and non-conductive atsome time during each half-cycle in response to the timing voltageV_(TIM).

FIG. 8 is a simplified timing diagram showing examples of thephase-control voltage V_(PC) generated by the dimmer switch 300, thetiming voltage V_(TIM), and the gate voltage V_(G) for driving the FETsQ210A, Q210B according to the third embodiment of the present invention.The phase-control voltage V_(PC) has a magnitude equal to approximatelythe magnitude of the AC line voltage V_(AC) of the AC power source 105at the beginning of each half-cycle during a conduction time T_(CON),and has a magnitude of approximately zero volts during the rest of thehalf-cycle, i.e., during a non-conduction time T_(NC). To generate thephase-control voltage V_(PC), the gate drive circuit 340 drives the gatevoltage V_(G) high towards the reference voltage V_(REF) at thebeginning of each half-cycle, such that the FETs Q210A, Q210B arerendered conductive (as shown at time t₁ in FIG. 8). At this time, thetiming circuit 330 begins generating the timing voltage V_(TIM), whichcomprises a ramp voltage that increases in magnitude with respect totime at a rate representative of the target intensity L_(TRGT) of theLED light source 104 (i.e., in response to the intensity adjustmentactuator 118). When the magnitude of the timing voltage V_(TIM) reachesa maximum timing voltage threshold V_(T-MAX) (e.g., approximately 7.5volts), the gate drive circuit 340 renders the FETs Q210A, Q210Bnon-conductive (as shown at time t₂ in FIG. 8). The rate of the timingvoltage V_(TIM) is inversely proportional to the target intensityL_(TRGT), i.e., the rate of the timing voltage V_(TIM) increases as thetarget intensity L_(TRGT) decreases, and decreases as the targetintensity L_(TRGT) increases. After the FETs Q210A, Q210B are renderednon-conductive, the gate drive circuit 340 will render the FETsconductive once again at the beginning of the next half-cycle (as shownat time t₃ in FIG. 8).

FIG. 9 is a simplified schematic diagram of the dimmer switch 300according to the third embodiment of the present invention. As shown inFIG. 9, the pass-transistor circuit 360 comprises an NPN bipolarjunction transistor Q362 having a collector coupled to receive therectifier voltage V_(RECT) through a resistor R364 (e.g., having aresistance of approximately 180Ω). The base of the transistor Q362 iscoupled to the rectifier voltage V_(RECT) through a resistor R365 (e.g.,having a resistance of approximately 470 kΩ), and to circuit commonthrough a zener diode Z366 (e.g., having a break-over voltage ofapproximately 15 volts). The pass-transistor circuit 360 furthercomprises a storage capacitor C368, which is able to charge through thetransistor Q362 and a diode D369 to a voltage equal to approximately thebreak-over voltage of the zener diode Z366 minus the base-emitter dropof the transistor Q362 and the forward drop of the diode D369. Thestorage capacitor C368 has, for example, a capacitance of approximately22 μF, and operates to maintain the reference voltage V_(REF) at anappropriate magnitude (e.g., at least approximately 12 volts) to controlthe FETs Q210A, Q210B to be conductive (i.e., when there isapproximately zero volts generated across the dimmer switch 100) as willbe described in greater detail below.

The snap-on circuit 370 is coupled to the storage capacitor Q368 andcomprises a PNP bipolar junction transistor Q372. The base of thetransistor Q372 is coupled to circuit common through the seriescombination of a resistor R374 (e.g., having a resistance ofapproximately 22 kΩ) and a zener diode Z376 (e.g., having a break-overvoltage of approximately 12 volts). The reference voltage V_(REF) isgenerated across a capacitor C378, which is coupled between thecollector of the transistor Q372 and circuit common and has, forexample, a capacitance of approximately 0.1 μF. The snap-on circuit 370operates such that the reference voltage V_(REF) is only provided acrossthe capacitor C378 when the magnitude of the voltage across the storagecapacitor C368 of the pass-transistor circuit 360 exceeds the break-overvoltage of the zener diode Z376 plus the emitter-base drop of thetransistor Q372.

The timing circuit 330 receives the reference voltage V_(REF) andgenerates the timing voltage V_(TIM) across a timing capacitor C332(e.g., having a capacitance of approximately 10 nF). The timing circuit330 includes a constant current source circuit for charging thecapacitor C332 at a constant rate to generate the timing voltageV_(TIM). The constant current source circuit comprises a PNP bipolarjunction transistor Q334 having an emitter coupled to the referencevoltage V_(REF) via a resistor R335 (e.g. having a resistance ofapproximately 180 kΩ). A voltage divider circuit comprising apotentiometer R336 and two resistors R338, R339 is coupled between thereference voltage V_(REF) and circuit common. For example, thepotentiometer R336 may have a resistance ranging from approximately 0 to500 kΩ, while the resistors R338, R339 may have resistances ofapproximately 100 kΩ and 82 kΩ, respectively. The junction of thepotentiometer R336 and the resistor R338 is coupled to the base of thetransistor Q334. The resistance of the potentiometer R336 varies inresponse to the intensity adjustment actuator 118 of the dimmer switch100, such that the magnitude of the voltage at the base of thetransistor Q334 is representative of the target intensity L_(TRGT). Whenthe potentiometer R336 is not presently being adjusted (i.e., is in asteady state condition), a constant voltage is generated across theresistor R335 and the emitter-base junction of the transistor Q334, suchthat the transistor Q334 conducts a constant current (having a magnitudedependent upon the magnitude of the voltage at the base of thetransistor Q334). Accordingly, the capacitor C332 charges at a ratedependent upon the target intensity L_(TRGT) thus generating the timingvoltage V_(TIM) (as shown in FIG. 8).

The gate drive circuit 340 renders the FETs Q210A, Q210B conductive atthe beginning of each half-cycle, and non-conductive at some time duringeach half-cycle in response to the timing voltage V_(TIM) from thetiming circuit 330. The gate drive circuit 340 comprises an NPN bipolarjunction transistor Q341 and a resistor R342, which is coupled betweenthe collector and base of the transistor Q341 and has a resistance of,for example, approximately 270 kΩ. A diode D343 is coupled between theemitter and the base of the transistor Q341. At the beginning of eachhalf-cycle, the resistor R342 conducts current into the base of thetransistor Q341. The transistor Q341 is thus rendered conductive and thereference voltage V_(REF) is coupled to the gates of the FETs Q210A,Q210B via the respective gate resistors R252, R254 to thus render theFETs conductive. As previously mentioned, the storage capacitor C368 ofthe voltage reference circuit 320 maintains the reference voltageV_(REF) at an appropriate magnitude (i.e., at least approximately 14.4volts) to maintain the FETs Q210A, Q210B conductive and the voltagedeveloped across the dimmer switch 300 is approximately zero volts.

The timing voltage V_(TIM) is coupled to the base of an NPN bipolarjunction transistor Q344 through a zener diode Z345 (e.g., having abreak-over voltage of approximately 6.8 volts). When the magnitude ofthe timing voltage V_(TIM) exceeds approximately the break-over voltageof the zener diode Z345 plus the base-emitter drop of the transistorQ344 (i.e., the maximum timing voltage threshold V_(T-MAX)), thetransistor Q344 is rendered conductive. Accordingly, the gate voltageV_(G) is pulled down towards circuit common through the diode D343 thusrendering the FETs Q210A, Q210B non-conductive.

The gate drive circuit 340 also comprises an NPN bipolar junctiontransistor Q346 coupled across the zener diode Z345. The base of thetransistor Q346 is coupled to the junction of two series-connectedresistors R347, R348 (e.g., having resistances of approximately 200 kΩand 10 kΩ respectively). The resistors R347, R348 form a voltage dividercoupled between the rectified voltage V_(RECT) and circuit common. Thebase of the transistor Q346 is also coupled to circuit common via acapacitor C349 (e.g., having a capacitance of approximately 10 nF). Whenthe FETs Q210A, Q210B are rendered non-conductive (in response to thetiming voltage V_(TIM) exceeding the maximum timing voltage thresholdV_(T-MAX)), the voltage developed across the dimmer switch 300 increasesto approximately the magnitude of the AC line voltage V_(AC) of the ACpower source 105. As a result, the voltage at the base of the transistorQ346 increases such that the transistor is rendered conductive.Accordingly, the magnitude of the timing voltage V_(TIM) is controlledto approximately zero volts and the transistor Q344 is maintainedconductive (thus keeping the FETs Q210A, Q210B non-conductive) until theend of the present half-cycle.

Near the end of the half-cycle, the magnitude of the AC line voltageV_(AC) of the AC power source 105 as well as the magnitude of voltage atthe base of the transistor Q346 decrease such that the transistor Q346is rendered non-conductive. Accordingly, the transistor Q344 is renderednon-conductive and the reference voltage V_(REF) is coupled to the gatesof the FETs Q210A, Q210B through the transistor Q341 and the respectivegate resistors R252, R254, thus rendering the FETs conductive. Inaddition, when the transistor Q346 is non-conductive, the timing voltageV_(TIM) of the timing circuit 330 may once again begin increasing inmagnitude with respect to time at the rate dependent upon the targetintensity L_(TRGT) (as shown in FIG. 8).

FIG. 10 is a simplified schematic diagram of a dimmer switch 400according to a fourth embodiment of the present invention. The dimmerswitch 400 of the fourth embodiment is very similar to the dimmer switch300 of the third embodiment. However, the dimmer switch 400 of thefourth embodiment comprises a voltage compensation circuit 480, whichreceives the rectified voltage V_(RECT) and adjusts the timing voltageV_(TIM) to account for changes and fluctuations in the AC line voltageV_(AC) of the AC power source 105 to avoid flickering of the intensityof the LED light source 104. The voltage compensation circuit 480comprises two resistors R482, R484, which are coupled in series betweenthe rectified voltage V_(RECT) and circuit common, and have, forexample, resistances of approximately 1 MΩ and 98 kΩ, respectively. Acapacitor C486 is coupled between the junction of the resistors R482,R484 and circuit common, and has, for example, a capacitance ofapproximately 0.22 μF. The capacitor C486 is coupled to the timingvoltage V_(TIM) through a resistor R488 (e.g., having a resistance ofapproximately 560 kΩ).

The voltage produced across the capacitor C486 is proportional to themagnitude of the AC line voltage V_(AC) of the AC power source 105 whenthe FETs Q210A, Q210B are non-conductive and the timing voltage V_(TIM)is increasing in magnitude with respect to time. When there are nochanges or fluctuations in the magnitude of the AC line voltage V_(AC)of the AC power source 105, the capacitor C486 charges to a steady-statevoltage. However, if the magnitude of the AC line voltage V_(AC) changeswhile the FETs Q210A, Q210B are non-conductive during a half-cycle(e.g., between times t₂ and t₃ in FIG. 8), the magnitude of the voltageacross the capacitor C486 will also change, thus resulting in a changein the timing voltage V_(TIM) when the FETs are conductive during thenext half-cycle (e.g., between times t₃ and t₄). For example, if themagnitude of the AC line voltage V_(AC) (and thus the magnitude of thevoltage across the capacitor C486) increases while the FETs Q210A, Q210Bare non-conductive during a half-cycle, the magnitude of the timingvoltage V_(TIM) will be greater while the FETs are conductive during thenext half-cycle, thus causing the FETs to be rendered non-conductiveearlier in the next half-cycle.

FIG. 11 is a simplified schematic diagram of a dimmer switch 500according to a fifth embodiment of the present invention. The dimmerswitch 500 comprises a mechanical air-gap switch S514 and two FETsQ510A, Q510B coupled in anti-series connection between the hot terminalH and the dimmed hot terminal DH for generating the phase-controlvoltage V_(PC). The dimmer switch 500 comprises an analog controlcircuit (e.g., a timing circuit 520) for generating a timing voltageV_(TIM) representative of the target intensity L_(TRGT) of the LED lightsource 104, and a gate drive circuit 530 for rendering the FETs Q510A,Q510B conductive and non-conductive in response to the timing voltageV_(TIM) to thus generate the phase-control voltage V_(PC). According tothe fifth embodiment of the present invention, the gate drive circuit530 is operable to generate two gate voltages V_(G1), V_(G2) forindependently controlling the respective FETs Q510A, Q510B on acomplementary basis. The FETs Q510A, Q510B are rendered conductive whenthe magnitudes of the respective gate voltages V_(G1), V_(G2) arecontrolled to a nominal gate voltage V_(N) (e.g., approximately 9 V) andare rendered non-conductive when the magnitudes of the respective gatevoltages V_(G1), V_(G2) are controlled to approximately zero volts. Thedimmer switch 500 further comprises an overcurrent protection circuit540 for rendering the FETs Q510A, Q510B non-conductive in the event ofan overcurrent condition in the FETs.

FIG. 12 is a simplified timing diagram showing examples of thephase-control voltage V_(PC) generated by the dimmer switch 500 and thegate voltages V_(G1), V_(G2) for driving the FETs Q510A, Q510B,respectively. According to the fifth embodiment of the presentinvention, the phase-control voltage V_(PC) comprises a forwardphase-control voltage. During the positive half-cycles, the first FETQ510A is rendered conductive and the second FET Q510B is renderednon-conductive when the first gate voltage V_(G1) increases fromapproximately zero volts to the nominal gate voltage V_(N) (as shown attime t₁), and the second gate voltage V_(G2) decreases from the nominalgate voltage V_(N) to approximately zero volts. At this time, the dimmerswitch 500 conducts the load current I_(LOAD) to the LED driver 102through the first FET Q510A and the body diode of the second FET Q510B.At the beginning of the negative half-cycles, the first FET Q510 remainsconductive. However, since the second FET Q510B is non-conductive andthe body diode of the second FET Q510B is reversed-biased, the dimmerswitch 500 does not conduct the load current I_(LOAD) at this time.

During the negative half-cycles, the first FET Q510A is renderednon-conductive and the second FET Q510B is rendered conductive when thefirst gate voltage V_(G1) decreases from the nominal gate voltage V_(N)to approximately zero volts and the second gate voltage V_(G2) increasesfrom approximately zero volts to the nominal gate voltage V_(N) (asshown at time t₂). At this time, the dimmer switch 500 conducts the loadcurrent I_(LOAD) to the LED driver 102 through the second FET Q510B andthe body diode of the first FET Q510A. At the beginning of the positivehalf-cycles, the second FET Q510B remains conductive, the first FETQ510A remains non-conductive, and the body diode of the first FET Q510Ais reversed-biased at this time, such that the dimmer switch 500 doesnot conduct the load current I_(LOAD) until the first FET Q510A isrendered conductive.

The timing circuit 520 is coupled in series between the hot terminal Hand the dimmed hot terminal DH and conducts a timing current I_(TIM)(i.e., a control current) through the LED driver 102 in order togenerate the timing voltage V_(TIM) across a capacitor C522 (e.g.,having a capacitance of approximately 0.1 μF). The capacitor C522 isoperable to charge from the AC power source 105 through resistors R524,R525 (e.g., having resistances of approximately 27 kΩ and 10 kΩ,respectively) and a potentiometer R526. The resistance of thepotentiometer R526 may range from, for example, approximately 0 kΩ to300 kΩ, and may be controlled by a user of the dimmer switch 500 (e.g.,by actuating the slider control) to adjust the target intensity L_(TRGT)of the LED light source 104. A calibration resistor R527 is coupled topotentiometer R526 for calibrating the range of the potentiometer, andhas a resistance of, for example, approximately 300 kΩ. Since thecapacitor C522 charges through the potentiometer R526, the rate at whichthe capacitor C522 charges and thus the magnitude of the timing voltageV_(TIM) are representative of the target intensity L_(TRGT) of the LEDlight source 104.

The drive circuit 530 comprises a diac 532 (e.g., having a break-overvoltage V_(BR) of approximately 32 volts) and two pulse transformers534A, 534B. The diac 532 is coupled in series with the primary windingsof the two pulse transformers 534A, 534B. The secondary windings of thepulse transformers 534A, 534B are coupled to respective capacitorsC535A, C535B via respective zener diodes Z536A, Z536B (which each have abreak-over voltage approximately equal to the nominal gate voltageV_(N), i.e., approximately 9 V). The capacitors C535A, C535B are coupledto the gates of the FETs Q510A, Q510B via gate resistors R538A, R538B,respectively (e.g., having resistances of approximately 47 kΩ). The gateresistors R538A, R538B may alternatively have different resistances inorder to change the duration of the switching times of the FETs Q510A,Q510B as is well known in the art.

When the magnitude of the timing voltage V_(TIM) exceeds approximatelythe break-over voltage V_(BR) of the diac 532, the diac conducts a pulseof current (i.e., a firing current I_(FIRE) as shown in FIG. 12) throughthe primary windings of the pulse transformers 534A, 534B causingsecondary voltages V_(SEC) (e.g., approximately 9V) to be generatedacross the secondary windings of the pulse transformers. During thepositive half-cycles, the capacitor C535A charges from the secondarywinding of the first pulse transformer 534A through the zener diodeZ536A to approximately the nominal gate voltage V_(N) (i.e.,approximately 9 volts). Accordingly, the first gate voltage V_(G1) isdriven high from approximately zero volts to the nominal gate voltageV_(N) rendering the first FET Q510A conductive (as shown at time t₁ inFIG. 12). At the beginning of the negative half-cycles, the first FETQ510A is conductive, while the second FET Q510B is non-conductive. Sincethe body diode of the second FET Q510B is reversed biased at this time,the dimmer switch 500 does not conduct the load current I_(LOAD).

During the negative half-cycles, the firing current I_(FIRE) has anegative magnitude, thus causing the secondary voltages V_(SEC) acrossthe secondary windings of the pulse transformers 534A, 534B to also havenegative magnitudes. Accordingly, the zener diode Z536A isreverse-biased during the negative half-cycles, causing the capacitorC535A to discharge through the zener diode Z536A, such that the voltageacross the capacitor C535A is driven to approximately zero volts. As aresult, the first gate voltage V_(G1) is driven low from the nominalgate voltage V_(N) to approximately zero volts rendering the first FETQ510A non-conductive (as shown at time t₂ in FIG. 12). In addition, thezener diode Z536B coupled to the secondary winding of the second pulsetransformer 534B is forward-biased in the negative half-cycles, suchthat the capacitor C535B charges to approximately the nominal gatevoltage V_(N) and the second FET Q510B is rendered conductive during thenegative half-cycles (as shown at time t₂ in FIG. 12). Accordingly, theFETs Q510A, Q510B are driven in a complementary manner, such that—at alltimes—at least one FET is conductive, while the other FET isnon-conductive. As a result, the FETs Q510A, Q510B are driven to beconductive for approximately the period T_(HC) of a half-cycle andnon-conductive for the period T_(HC) of a half-cycle.

The timing circuit 520 also comprises a diac 528 (e.g., having abreak-over voltage of approximately 64V) coupled to the potentiometerR526. The diac 528 provides voltage compensation by adjusting thevoltage provided to the potentiometer R526 to compensate for variationsin the AC line voltage V_(AC) provided by the AC power source 105. Thediac 528 has a negative impedance transfer function, such that thevoltage across the diac increases as the current through the diacdecreases. Thus, as the voltage across the dimmer switch 500 (i.e.,between the hot terminal H and the dimmed hot terminal DH) decreases,the current through the resistor R524 and the diac 528 decreases. As aresult, the voltage across the diac 528 increases, thus causing thecurrent flowing through the potentiometer R526 to increase and thefiring capacitor C522 to charge at a faster rate. This results in anincreased conduction time T_(CON) of the FETs Q510A, Q510B during thepresent half-cycle to compensate for the decreased voltage across thedimmer switch 500, thereby maintaining the intensity of the LED lightsource 104 constant.

The drive circuit 530 is characterized as having inherent shorted-FETprotection. In the event that one of the FETs Q510A, Q510B failsshorted, the drive circuit 530 is operable to drive the other,non-shorted FET into full conduction, such that the load currentI_(LOAD) is not asymmetric. Asymmetric current can cause some types oflighting loads to overheat. For example, if the second FET Q510B failsshorted, the full AC waveform will be provided to the LED driver 102during the negative half-cycles. Since there will be approximately zerovolts produced across the dimmer switch 500 during the negativehalf-cycles when second FET Q510B is shorted, the capacitor C522 of thetiming circuit 520 will not charge, the diac 532 of the drive circuit330 will not conduct the pulse of the firing current I_(FIRE), and thevoltage across the capacitor C535A will not be driven to zero volts torender the first FET Q510A non-conductive during the negativehalf-cycles. Accordingly, the first FET Q510A will remain conductiveduring both half-cycles and the load current I_(LOAD) will besubstantially symmetric. The second FET Q510B is controlled to beconductive in a similar manner if the first FET Q510A has failedshorted.

The overcurrent protection circuit 540 comprises a sense resistor R542(e.g., having a resistance of approximately 0.015Ω). The sense resistorR542 is coupled between the sources of the FETs Q510A, Q510B, such thata voltage representative of the magnitude of the load current I_(LOAD)is generated across the sense resistor. The voltage generated across thesense resistor R542 is provided to the base of a first NPN bipolarjunction transistor (BJT) Q544. The first transistor Q544 is coupledacross the capacitor C535A and operates to protect the first FET Q510Ain the event of an overcurrent condition during the positivehalf-cycles. When the magnitude of the load current I_(LOAD) exceeds apredetermined current limit (e.g., approximately 46.6 amps) such thatthe voltage generated across the sense resistor R542 exceeds the ratedbase-emitter voltage (e.g., approximately 0.7 volts) of the firsttransistor Q544, the first transistor is rendered conductive.Accordingly, the first transistor Q544 pulls the first gate voltageV_(G1) at the gate of the first FET Q510A down towards zero volts, thusrendering the first FET non-conductive. The overcurrent protectioncircuit 540 further comprises a second NPN bipolar junction transistorQ546, which is coupled across the capacitor C535B and operates toprotect the second FET Q510B during the negative half-cycles. When themagnitude of the load current I_(LOAD) exceeds the predetermined currentlimit, the second transistor Q546 is rendered conductive, thus pullingthe second gate voltage V_(G2) at the gate of the second FET Q510B downtowards zero volts and rendering the second FET non-conductive.

FIG. 13 is a simplified schematic diagram of a dimmer switch 600according to a sixth embodiment of the present invention. The dimmerswitch 600 comprises a drive limit circuit 650, which is coupled inseries with the diac 532 and the primary windings of the two pulsetransformers 534A, 534B of the drive circuit 530. The drive limitcircuit 650 operates to limit the number of times that the drive circuit530 attempts to render the FETs Q510A, Q510B conductive during aspecific half-cycle. For example, if the overcurrent protection circuit540 renders one of the FETs Q510A, Q510B non-conductive, the drive limitcircuit 650 prevents the drive circuit 530 from attempting to render therespective FET conductive again during the present half-cycle.

When the diac 532 fires each half-cycle, the drive limit circuit 650conducts the firing current I_(FIRE) and generates an offset voltageV_(OFFSET) across a capacitor C652A during the positive half-cycles anda capacitor C652B during the negative half-cycles. The capacitor C452Acharges through a diode D654A during the positive half-cycles, and thecapacitor C452B charges through a diode D654B during the negativehalf-cycles. For example, the capacitors C652A, C652B may havecapacitances of approximately 0.1 μF. Discharge resistors R656A, R656Bare coupled in parallel with the capacitors C652A, C652B, respectively,and each have a resistance of, for example, approximately 33 kΩ. Thedrive limit circuit 450 further comprises two zener diodes Z658A, Z658Bcoupled in anti-series connection and each having the same break-overvoltage V_(Z) (e.g., approximately 40V). The zener diodes Z658A, Z658Bare coupled to the timing circuit 520 to limit the magnitude of thetiming voltage V_(TIM) to a clamp voltage V_(CLAMP), i.e., approximatelythe break-over voltage V_(Z), in both half-cycles.

At the beginning of a positive half-cycle, the capacitor C652A of thedrive limit circuit 540 has no charge, and thus, no voltage is developedacross the capacitor. The timing voltage signal V_(TIM) increases untilthe magnitude of the timing voltage V_(TIM) exceeds approximately thebreak-over voltage V_(BR) of the diac 532. When the diac 532 fires, thediode D654A and the capacitor C652A conduct pulse of the firing currentI_(FIRE) and the offset voltage V_(OFFSET) (e.g., approximately 12volts) is developed across the capacitor C652A. After the diac 532 hasfinished conducting the firing current I_(FIRE), the voltage across thecapacitor C522 decreases by approximately a break-back voltage (e.g.,approximately 10 volts) of the diac 532 to a predetermined voltage V_(P)(e.g., approximately 22 volts). If the overcurrent protection circuit540 renders one of the FETs Q510A, Q510B non-conductive, the timingvoltage signal V_(TIM) will begin to increase again. The magnitude ofthe timing voltage V_(TIM) must exceed approximately the break-overvoltage V_(BR) of the diac 532 plus the offset voltage V_(OFFSET) acrossthe capacitor C652A (i.e., approximately 44 volts) in order for the diac532 to conduct the pulse of the firing current I_(FIRE) once again.However, because the zener diode Z658A limits the timing voltage V_(TIM)to the break-over voltage V_(Z) (i.e., approximately 40 volts), thetiming voltage V_(TIM) is prevented from exceeding the voltage thresholdV_(TH). Accordingly, the drive circuit 530 is prevented from repeatedlyattempting to render the FETs Q510A, Q510B conductive during eachhalf-cycle in the event of an overcurrent condition.

The timing voltage V_(TIM) is prevented from exceeding the voltagethreshold V_(TH) until the voltage ΔV across the capacitor C652A decaysto approximately the break-over voltage V_(Z) of the zener diode Z658Aminus the break-over voltage V_(BR) of the diac 532. The capacitor C652Adischarges slowly through the discharge resistor R656A, such that thetime required for the voltage ΔV across the capacitor C652A to decay toapproximately the break-over voltage V_(Z) of the zener diode Z658Aminus the break-over voltage V_(BR) of the diac 532 is long enough suchthat the drive circuit 530 only attempts to render the FETs Q510A, Q510Bconductive once during each half-cycle. The voltage across the capacitorC652A decays to substantially zero volts during the negative half-cyclesuch that the voltage across the capacitor C652A is substantially zerovolts at the beginning of the next positive half-cycle. The capacitorC652B, the diode D654B, the discharge resistor R656B, and the zenerdiode Z658B of the drive limit circuit 650 operate in a similar fashionduring the negative half-cycles. An example of the drive limit circuit650 is described in greater detail in commonly-assigned U.S. Pat. No.7,570,031, issued Aug. 4, 2009, entitled METHOD AND APPARATUS FORPREVENTING MULTIPLE ATTEMPTED FIRINGS OF A SEMICONDUCTOR SWITCH IN ALOAD CONTROL DEVICE, the entire disclosure of which is herebyincorporated by reference.

FIG. 14 is a simplified schematic diagram of a dimmer switch 700according to a seventh embodiment of the present invention. The dimmerswitch 700 comprises a drive circuit 730 that includes a single pulsetransformer 734. The pulse transformer 734 has a single primary windingand secondary winding having a tap connection 734′. The diac 532 iscoupled in series with the single primary winding of the pulsetransformer 734. The series combination of the zener diode Z536A and thecapacitor C535A is coupled between one end of the secondary winding andthe tap connection 734′ of the pulse transformer 734. The seriescombination of the diode Z536B and the capacitor C535B is coupledbetween the other end of the secondary winding and the tap connection734′ of the pulse transformer 734. The drive circuit 730 of the seventhembodiment operates to render the FETs Q510A, Q510B conductive andnon-conductive in the same manner as the drive circuit 530 of the fifthembodiment.

FIG. 15 is a simplified schematic diagram of a dimmer switch 800according to an eighth embodiment of the present invention. The dimmerswitch 800 comprises a mechanical air-gap switch S814 and two FETsQ810A, Q810B coupled in anti-series connection between the hot terminalH and the dimmed hot terminal DH for control of the amount of powerdelivered to the connected LED driver 102. As in the fifth, sixth, andseventh embodiments, the FETs Q810A, Q810B have control inputs (i.e.,gates) that receive respective gate voltages V_(G1), V_(G2) forrendering the FETs conductive and non-conductive. The LED light source104 is off when the switch S814 is open, and is on when the switch isclosed. The dimmer switch 800 comprises a control circuit that includesa timing circuit 820 and a power supply 880 and is operable to conduct acontrol current I_(CNTL) through the LED driver 102. The timing circuit820 conducts a timing current I_(TIM) in order to generate a timingvoltage V_(TIM) (as in the fifth embodiment). The dimmer switch 800further comprises a drive circuit 830 for rendering the FETs 810A, Q810Bconductive and non-conductive in response to the timing voltage V_(TIM)and an overcurrent protection circuit 860 for rendering the FETs 810A,Q810B non-conductive in response to an overcurrent condition through theFETs.

The power supply 880 generates a DC supply voltage V_(S) (e.g.,approximately 14.4 volts) for powering the drive circuit 830 and theovercurrent protection circuit 860. The power supply 880 conducts acharging current I_(CHRG) through the LED driver 102 when the dimmerswitch 800 is not conducting the load current I_(LOAD) to the LED driverand the magnitude of the voltage developed across the dimmer switch isapproximately equal to the magnitude of the AC line voltage V_(AC). Thecontrol current I_(CNTL) conducted through the LED driver 102 isapproximately equal to the sum of the timing current I_(TIM) of thetiming circuit 820 and the charging current I_(CHRG) of the power supply880.

The power supply 880 comprises a diode D881 coupled to the hot terminalH (via the switch S814), such that the power supply 880 only chargesduring the positive half-cycles of the AC power source 105. The powersupply 880 includes a pass-transistor circuit that operates to generatethe supply voltage V_(S) across a capacitor C882 (e.g., having acapacitance of approximately 10 μF). The pass-transistor circuitcomprises an NPN bipolar junction transistor Q883, a resistor R884(e.g., having a resistance of approximately 220Ω), a resistor R885(e.g., having a resistance of approximately 470 kΩ), and a zener diodeZ886. The capacitor C882 is coupled to the emitter of the transistorQ883, such that the capacitor is able to charge through the transistor.The zener diode Z886 is coupled to the base of the transistor Q883 andhas a break-over voltage of, for example, approximately 15V, such thatthe capacitor C882 is able to charge to a voltage equal to approximatelythe break-over voltage minus the base-emitter drop of the transistor.

The power supply 880 further comprises snap-on circuit including a PNPbipolar junction transistor Q887, a resistor R888 (e.g., having aresistance of approximately 22 kΩ), and a zener diode Z889. The resistorR888 and the zener diode Z889 are coupled in series with the base of thetransistor Q887, and the collector of the transistor Q887 is coupled toa capacitor C890. The zener diode Z889 has a break-over voltage of, forexample, approximately 12 V, such that the voltage across the capacitorC882 is coupled across the capacitor C890 when the magnitude of thevoltage across the capacitor C882 exceeds approximately the break-overvoltage of the zener diode Z889 plus the emitter-base drop of thetransistor Q887. When the magnitude of the voltage across the capacitorC882 drops below approximately the break-over voltage of the zener diodeZ889 plus the emitter-base drop of the transistor Q887, the voltageacross the capacitor C882 is disconnected from the capacitor C890, suchthat the supply voltage V_(S) will drop to approximately circuit common(i.e., approximately zero volts).

The timing circuit 820 conducts the timing current I_(TN) and generatesthe timing voltage V_(TIM) across a capacitor C822 (e.g., having acapacitance of approximately 0.047 μF). The capacitor C822 charges fromthe AC power source 105 through resistors R824, R825 (e.g., havingresistances of approximately 27 kΩ and 10 kΩ, respectively) and apotentiometer R826 (e.g., having a resistance ranging from approximately0 kΩ to 300 kΩ). A calibration potentiometer R827 is coupled across thepotentiometer R826 and has, for example, a resistance ranging fromapproximately 0 to 500 kΩ. The timing circuit 820 further comprises adiac 828, which has a break-over voltage of, for example, approximately64V, and operates to provide voltage compensation for the timing circuit(in a similar manner as the diac 528 of the timing circuit 520 of thefifth embodiment).

The drive circuit 830 generates the gate voltages V_(G1), V_(G2) forrendering the FETs Q810A, Q810B conductive and non-conductive on acomplementary basis in response to the timing voltage V_(TIM) of thetiming circuit 820. The drive circuit 830 comprises a diac 832 (e.g.,having a break-over voltage of approximately 32 volts), a resistor R834(e.g., having a resistance of approximately 680Ω), and two optocouplersU835A, U835B. When the magnitude of the timing voltage V_(TIM) exceedsapproximately the break-over voltage of the diac 832, the diac conductsa firing current I_(FIRE) through the input photodiode of the firstoptocoupler U835A during the positive half-cycles, and through the inputphotodiode of the second optocoupler U835B during the negativehalf-cycles. Accordingly, the output phototransistor of the firstoptocoupler U835A is rendered conductive during the positivehalf-cycles, and the output phototransistor of the second optocouplerU835B is rendered conductive during the negative half-cycles. The outputphototransistors of the optocouplers U835A, U835B are between the supplyvoltage V_(S) and circuit common through respective resistors R836,R838, which each have resistances of, for example, approximately 4.7 kΩ.

The output phototransistors of the optocouplers U835A, U835B are alsocoupled to set-reset (SR) latches U840A, U840B, U840C, U840D, whichoperate to generate the gate voltages V_(G1), V_(G2) and to thus renderthe FETs Q810A, Q810B conductive and non-conductive on the complementarybasis. For example, the SR latches U840A, U840B, U840C, U840D may beimplemented as part of a single integrated circuit (IC), which may bepowered by the supply voltage V_(S). As shown in FIG. 15, the outputphototransistor of the first optocoupler U835A is coupled to the setinput of the first SR latch U840A and to the reset input of the secondSR latch U840B. The output phototransistor of the second optocouplerU835B is coupled to the set input of the second SR latch U840B and tothe reset input of the first SR latch U840A. The output of the first SRlatch U840A is coupled to the gate of the first FET Q810A and the outputof the second SR latch U840B is coupled to the gate of the second FETQ810B through respective resistors R842, R852, which each have aresistance of, for example, approximately 47 kΩ.

When the output phototransistor of the first optocoupler U835A isrendered conductive during the positive half-cycles, the output of thefirst SR latch U840A is driven high towards the supply voltage V_(S)(thus rendering the first FET Q810A conductive), while the output of thesecond SR latch U840B is driven low towards circuit common (thusrendering the second FET Q810B non-conductive). Similarly, when theoutput phototransistor of the second optocoupler U835B is renderedconductive during the negative half-cycles, the output of the second SRlatch U840B is driven high towards the supply voltage V_(S) (thusrendering the second FET Q810B conductive), while the output of thefirst SR latch U840A is driven low towards circuit common (thusrendering the first FET Q810A non-conductive). Since the set input ofthe first SR latch U840A is coupled to the reset input of the second SRlatch U840B, and the set input of the second SR latch is coupled to thereset input of the first SR latch, the FETs Q810A, Q810B are driven in acomplementary manner (as in the fifth embodiment), such that one of theFETs is conductive, while the other FET is non-conductive.

The overcurrent protection circuit 860 is coupled to the set inputs ofthe third and fourth SR latches U840C, U840D for rendering the FETsQ810A, Q810B non-conductive in the event of an overcurrent conditionthrough the FETs. The output of the third SR latch U840C is coupled tothe base of an NPN bipolar junction transistor Q844 via a resistor R846(e.g., having a resistance of approximately 18 kΩ). The collector of thetransistor Q844 is coupled to the gate of the first FET Q810A via aresistor R848 (e.g., having a resistance of approximately 330Ω). Thedrive circuit 830 comprises a similar circuit for coupling the output ofthe fourth SR latch U840D to the gate of the second FET Q810B.

The overcurrent protection circuit 860 comprises a sense resistor R870(e.g., having a resistance of approximately 0.015Ω). The sense resistorR870 is coupled in series between the FETs Q810A, Q810B, and circuitcommon is referenced to one side of the sense resistor (as shown in FIG.10), such that the magnitude of the voltage generated across the senseresistor is proportional to the magnitude of the load current I_(LOAD).The sense resistor R870 is coupled to the base of an NPN bipolarjunction transistor Q861 via a resistor R862 (e.g., having a resistanceof approximately 2.2 kΩ). A resistor R863 is coupled between the baseand the emitter of the transistor Q861 and has a resistance of, forexample, approximately 4.7 kΩ. The emitter of the transistor Q861 iscoupled to circuit common and the collector is coupled to the supplyvoltage V_(S) via two resistors R864, R865 (e.g., having resistances ofapproximately 18 kΩ and 4.7 kΩ, respectively). The junction of theresistors R864, R865 is coupled to the base of a PNP bipolar junctiontransistor Q866. The emitter of the transistor Q866 is coupled to thesupply voltage V_(S) and the collector is coupled to circuit commonthrough a resistor R867 (e.g., having a resistance of approximately510Ω). The collector of the transistor Q866 is coupled to the set inputof the third SR latch U840C for rendering the first FET Q810Anon-conductive in the event of overcurrent conditions during thepositive half-cycles. The overcurrent protection circuit 860 comprises asimilar circuit (including transistors Q871, Q876, and resistors R872,R873, R874, R875, R877) for rendering the second FET Q810Bnon-conductive in the event of overcurrent conditions during thenegative half-cycles.

In the event of an overcurrent condition during a positive half-cycle,the overcurrent protection circuit 860 drives the set input of the thirdSR latch U840C high towards the supply voltage V_(S). Thus, thetransistor Q844 is rendered conductive pulling the gate voltage V_(G1)down towards circuit common and rendering the first FET Q810Anon-conductive. The output phototransistor of the second optocouplerU835B is coupled to the reset input of the third SR latch U840C, suchthat the overcurrent protection is reset during the next half-cycle(i.e., the negative half-cycle). Specifically, when the outputphototransistor of the second optocoupler U835B is rendered conductiveduring the negative half-cycles, the reset input of the third SR latchU840C latch is driven high towards the supply voltage V_(S), thusrendering the transistor Q844 non-conductive and allowing the first SRlatch U840A to control the first FET Q810A. Similarly, the overcurrentprotection circuit 860 drives the set input of the fourth SR latch U840Dhigh towards the supply voltage V_(S), thus rendering the second FETQ810B non-conductive in the event of an overcurrent condition during anegative half-cycle. The reset input of the fourth SR latch U840D isdriven high when the output phototransistor of the first optocouplerU835A is rendered conductive during the positive half-cycles, thusallowing the second SR latch U840B to once again control the second FETQ810B.

FIG. 16 is a simplified schematic diagram of a dimmer switch 900according to a ninth embodiment of the present invention. The dimmerswitch 900 of the ninth embodiment is similar to the dimmer switch 100of the first embodiment (as shown in FIG. 4). However, the dimmer switch900 of the ninth embodiment comprises a digital control circuit 915having a microprocessor 930 for generating a drive voltage V_(DR) (whichis the same as the drive voltage V_(DR) of the first embodiment shown inFIGS. 3A and 3B). Alternatively, the microprocessor 930 may beimplemented as a microcontroller, a programmable logic device (PLD), anapplication specific integrated circuit (ASIC), a field-programmablegate array (FPGA), or any suitable controller or processing device. Asshown in FIG. 16, the bidirectional semiconductor switch 110 isimplemented as the triac 110′. Alternatively, the bidirectionalsemiconductor switch 110 of the dimmer switch 900 could be implementedas two FETs in anti-series connection that are simultaneously controlledto be conductive and non-conductive (i.e., in a similar manner as theFETs Q210A, Q210B of the dimmer switch 200 of the second embodiment).

The digital control circuit 915 also comprises a power supply 920operable to conduct a charging current I_(CHRG) through the LED driver102 in order to generate a DC supply voltage V_(CC). For example, thepower supply 920 may comprise a pass-transistor circuit (as in thedimmer switch 100 of the first embodiment shown in FIG. 4) or anysuitable power supply that does not draw a large charging currentthrough the LED driver 102. The digital control circuit 915 comprises avoltage divider having two resistors R934, R935 for generating a scaledvoltage V_(SCALED) having a magnitude suitable to be provided to themicroprocessor 930. The scaled voltage V_(SCALED) is representative ofthe voltage developed across the bidirectional semiconductor switch 110.The microprocessor 930 may have an analog-to-digital converter (ADC) forsampling the scaled voltage V_(SCALED), such that the microprocessor 930is operable to determine the zero-crossings of the phase control voltageV_(PC) in response to the voltage developed across the bidirectionalsemiconductor switch 110.

The digital control circuit 915 further comprises a toggle tactileswitch S_(TOGGLE), a raise tactile switch S_(RAISE), and a lower tactileswitch S_(LOWER) for receiving user inputs. The toggle tactile switchS_(TOGGLE) may be mechanically coupled to a toggle actuator or pushbutton. The raise and lower switches S_(RAISE), S_(LOWER) may bemechanically coupled to, for example, separate raise and lower buttons,respectively, or to a rocker switch having an upper portion and a lowerportion. The toggle switch S_(TOGGLE) is coupled in series with aresistor R936 between the supply voltage V_(CC) and circuit common, andgenerates a toggle control signal V_(TOGGLE). The raise switch S_(RAISE)is coupled in series with a resistor R938 between the supply voltageV_(CC) and circuit common, and generates a raise control signalV_(RAISE). The lower switch S_(LOWER) is coupled in series with aresistor R938 between the supply voltage V_(CC) and circuit common, andgenerates a lower control signal V_(LOWER). The toggle control signalV_(TOGGLE), the raise control signal V_(RAISE), and the lower controlsignal V_(LOWER) are received by the microprocessor 930. Themicroprocessor 930 is operable to toggle the LED light source 104 on andoff in response to subsequent actuations of the toggle switchS_(TOGGLE). The microprocessor 930 is operable to increase the targetintensity L_(TRGT) of the LED light source 104 in response to actuationsof the raise switch S_(RAISE) and to decrease the target intensityL_(TRGT) in response to actuations of the lower switch S_(LOWER).Alternatively, the digital control circuit 915 could comprise apotentiometer for generating a DC voltage that is representative of thedesired intensity of the LED light source 104 and varies, for example,in magnitude in response to the position of an intensity adjustmentactuator of the dimmer switch 900 (i.e., similar to the potentiometerR144 and the intensity adjustment actuator 118 of the dimmer switch 100of the first embodiment).

In addition, the microprocessor 930 of the dimmer switch 900 mayalternatively be operable to receive a digital message from a wired orwireless signal receiver. For example, the digital control circuit 915of the dimmer switch 900 may comprise a radio-frequency (RF) transceiver(not shown) and an antenna (not shown) for transmitting and receiving RFsignals. The microprocessor 930 may be operable to control thebidirectional semiconductor switch 110 in response to the digitalmessages received via the RF signals. Alternatively, the dimmer switch900 may simply comprise an RF receiver or an RF transmitter for onlyreceiving or transmitting RF signals, respectively. Examples of RF loadcontrol devices and antennas for wall-mounted load control devices aredescribed in greater detail in commonly-assigned U.S. Pat. No.5,982,103, issued Nov. 9, 1999, and U.S. Pat. No. 7,362,285, issued Apr.22, 2008, both entitled COMPACT RADIO FREQUENCY TRANSMITTING ANDRECEIVING ANTENNA AND CONTROL DEVICE EMPLOYING SAME, the entiredisclosures of which are hereby incorporated by reference.

FIG. 17 is a simplified flowchart of a switch procedure 1000 executed bythe microprocessor 930 in response to an actuation of one of the raiseswitch S_(RAISE) or the lower switch S_(LOWER) at step 1010 (i.e., ifeither of the raise control signal V_(RAISE) and the lower controlsignal V_(LOWER) are pulled down to circuit common). If the raise switchS_(RAISE) is actuated at step 1012, the microprocessor 930 increases thetarget intensity L_(TRGT) of the LED light source 104 at step 1014 bydecreasing a firing time T_(FIRE) (which is approximately equal to thenon-conduction time T_(NC) shown in FIGS. 3A and 3B). If the lowerswitch S_(LOWER) is actuated at step 1016, the microprocessor 930decreases the target intensity L_(TRGT) of the LED light source 104 byincreasing the firing time T_(FIRE) at step 1018, before the buttonprocedure 1000 exits.

FIG. 18 is a simplified flowchart of a control procedure 1100periodically executed by the microprocessor 930 (e.g., every 100 μsec)to sample the scaled voltage V_(SCALED) and generate the drive voltageV_(DR). First, the microprocessor 930 samples the scaled voltageV_(SCALED) using the ADC at step 1110. At step 1112, the microprocessor930 determines if the scaled voltage V_(SCALED) is increasing inmagnitude and if the present sample is greater than the previous samplein order to detect a positive-going transition of the scaled voltageV_(SCALED) across a zero-crossing threshold. If the microprocessor 930detects a positive-going transition across the zero-crossing thresholdat step 1112 and a RESET flag is set at step 1114, the microprocessor930 clears the RESET flag at step 1116. The microprocessor 930 theninitializes a timer to zero and starts the timer increasing in valuewith respect to time at step 1118, before the control procedure 1100exits. If the RESET flag is not set at step 1114, the microprocessor 930does not restart the timer at step 1118.

If the timer is equal to the firing time T_(FIRE) at step 1120, themicroprocessor 930 drives the drive voltage V_(DR) low to approximatelycircuit common to render the bidirectional semiconductor switch 110conductive at step 1122, and the control procedure 1100 exits. If thetime is equal to a total time T_(TOTAL) at step 1124, the microprocessor930 drives the drive voltage V_(DR) high to approximately the supplyvoltage V_(CC) to render the bidirectional semiconductor switch 110non-conductive at step 1126. The total time T_(TOTAL) may be equal tothe fixed amount of time T_(TIM) that the timing circuit 130 generatesthe timing voltage V_(TIM) in the dimmer switch 100 of the firstembodiment (i.e., approximately 7.5 msec). At step 1128, themicroprocessor 930 sets the RESET flag at step 1128, and the controlprocedure 1100 exits. The RESET flag allows the microprocessor 930 toensure that the timer is not restarted until after the total timeT_(TOTAL).

While the present invention has been described with reference to thehigh-efficiency lighting load 101 having the LED driver 102 forcontrolling the intensity of the LED light source 104, the dimmerswitches 100, 200, 300, 400, 500, 600, 700, 800, 900 could be used tocontrol the amount of power delivered to other types of lighting loads(such as incandescent lamps, halogen lamps, magnetic low-voltage lamps,electronic low-voltage lamps), other types of electrical loads (such asmotor and fan loads), and other types of load regulation devices (suchas electronic dimming ballasts for fluorescent lamps).

This application is related to commonly-assigned U.S. PatentApplication, Attorney Docket No. P/10-1508 (10-21992-P2), filed Nov. 23,2010, entitled TWO-WIRE ANALOG FET-BASED DIMMER SWITCH, the entiredisclosure of which is hereby incorporated by reference.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein, but only by the appended claims.

What is claimed is:
 1. A load control device for controlling powerdelivered from an AC power source to an electrical load, the loadcontrol device comprising: a thyristor adapted to be electricallycoupled between the AC power source and the electrical load, thethyristor having first and second main terminals through which a loadcurrent can be conducted to energize the electrical load and a gateterminal through which a gate current can be conducted to render thethyristor conductive; a gate coupling circuit electrically coupled toconduct the gate current through the gate terminal of the thyristor; acontrol circuit electrically coupled to control the gate couplingcircuit, the control circuit configured to determine a zero-crossingtime of a present half-cycle of the AC power source in response to avoltage developed across the thyristor; and a power supply electricallycoupled to conduct a charging current through the electrical load togenerate a supply voltage; wherein the control circuit is configured tocontrol the gate coupling circuit to be conductive to conduct the gatecurrent through the gate terminal of the thyristor to render thethyristor conductive after a first, variable amount of time has elapsedsince the zero-crossing time of the present half-cycle, the controlcircuit configured to maintain the thyristor conductive independent ofthe magnitude of the load current conducted through the thyristor duringthe present half-cycle, the control circuit further configured toprevent the gate coupling circuit from conducting the gate currentthrough the gate terminal of the thyristor after a second, fixed amountof time has elapsed since the zero-crossing time of the presenthalf-cycle to allow the thyristor to become non-conductive, the second,fixed amount of time being less than a length of the present half-cycle,the control circuit configured to vary the first, variable amount oftime in accordance with a determinate amount of power to be delivered tothe electrical load in order to cause the amount of power delivered tothe electrical load to be the determinate amount.
 2. The load controldevice of claim 1, wherein the control circuit comprises: a timingcircuit configured to generate a timing signal from the supply voltage,the timing circuit being configured to start to generate the timingsignal near the zero-crossing time of the present half-cycle, and tocause the timing signal to increase in magnitude with respect to time;and a drive circuit arranged to receive the timing signal, the drivecircuit being configured to render the thyristor conductive in eachhalf-cycle in response to the timing signal reaching a selectedthreshold; wherein the timing circuit is further configured to continuegenerating the timing signal after the thyristor is rendered conductivein each half-cycle, resulting in the control circuit maintaining thethyristor conductive independent of the magnitude of the load currentconducted through the thyristor.
 3. The load control device of claim 2,wherein the timing circuit is configured to stop generating the timingsignal after the second, fixed amount of time has elapsed since thezero-crossing time of the present half-cycle.
 4. The load control deviceof claim 3, wherein the drive circuit is configured to terminate thedrive signal when the timing circuit stops generating the timing signalso as to render the gate coupling circuit non-conductive and prevent thegate terminal of the thyristor from conducting the gate current andthereby allow the thyristor to commutate off.
 5. The load control deviceof claim 3, wherein the timing circuit is configured to increase themagnitude of the timing signal at a constant rate and the length of thefirst, variable amount of time is not dependent upon the length of thesecond, fixed amount of time.
 6. The load control device of claim 1,wherein the control circuit is configured to generate a drive signal andapply the drive signal to the gate coupling circuit, the control circuitconfigured to adjust the magnitude of the drive signal to a firstmagnitude to render the gate coupling circuit conductive after thefirst, variable amount of time has elapsed since the zero-crossing time,maintain the magnitude of the drive signal at the first magnitude afterthe gate coupling circuit is rendered conductive, and adjust themagnitude of the drive signal to a second magnitude to prevent the gatecoupling circuit from conducting the gate current through the gateterminal of the thyristor after the second, fixed amount of time haselapsed since the zero-crossing time.
 7. The load control device ofclaim 6, wherein a dead time exists between the time at which thecontrol circuit adjusts the drive signal to the second magnitude duringa first half-cycle and the time at which the control circuit determinesa subsequent zero-crossing time in a second half-cycle immediatelyfollowing the first half-cycle.
 8. The load control device of claim 1,wherein the power supply is configured to generate the supply voltageacross a storage capacitor when the thyristor is non-conductive, thegate coupling circuit comprises an opto-coupler having an inputphotodiode electrically coupled to receive the drive signal from thecontrol circuit and an output photodiode electrically coupled to thegate terminal of the thyristor, the input photodiode configured toconduct a drive current from the storage capacitor of the power supplyin order to render the thyristor conductive and maintain the thyristorconductive in response to the gate drive signal.
 9. The load controldevice of claim 1, wherein the thyristor comprises a triac.
 10. The loadcontrol device of claim 1, wherein the electrical load comprises ahigh-efficiency lighting load and the load control device is a dimmerswitch.
 11. The load control device of claim 10, wherein thehigh-efficiency lighting load is a light-emitting diode light source ora compact fluorescent lamp.
 12. The load control device of claim 1,wherein the control circuit comprises a microprocessor powered by thesupply voltage.
 13. The load control device of claim 1, furthercomprising: an RF receiver configured to receive an RF signal; whereinthe control circuit is configured to determine the determinate amount ofpower to be delivered to the electrical load from the RF signal.
 14. Theload control device of claim 1, wherein the electrical load is alight-emitting diode light source or a compact fluorescent lamp.
 15. Theload control device of claim 1, wherein the control circuit isconfigured render the gate coupling circuit non-conductive to preventthe gate coupling circuit from conducting
 15. The load control device ofclaim 1, wherein the control circuit is configured render the gatecoupling circuit non-conductive to prevent the gate coupling circuitfrom conducting the gate current through the gate terminal of thethyristor after the second, fixed amount of time has elapsed since thezero-crossing time of the present half-cycle.
 16. The load controldevice of claim 1, wherein the control circuit is further configured tocontrol the second, fixed amount of time to be approximately equalduring each half-cycle of the AC power source.
 17. The load controldevice of claim 1, wherein the control circuit and the gate couplingcircuit are arranged in such a way as to enable the thyristor to conductthe load current to the electrical load and to conduct the load currentfrom the electrical load during a single half-cycle of the AC powersource.
 18. A load control device for controlling power delivered froman AC power source to an electrical load, the load control devicecomprising: a thyristor adapted to be electrically coupled between theAC power source and the electrical load, the thyristor having first andsecond main terminals through which a load current can be conducted toenergize the electrical load and a gate terminal through which a gatecurrent can be conducted to render the thyristor conductive; a gatecoupling circuit electrically coupled to conduct the gate currentthrough the gate terminal of the thyristor; a control circuitelectrically coupled to control the gate coupling circuit, the controlcircuit configured to control the gate coupling circuit to conduct thegate current through the gate terminal of the thyristor to render thethyristor conductive at a firing time during each half-cycle of the ACpower source; and a power supply electrically coupled to conduct acharging current through the electrical load to generate a supplyvoltage; wherein the control circuit is configured to maintain the gatecoupling circuit conductive after the firing time, thereby enabling thethyristor both to conduct the load current to the electrical load and toconduct the load current from the electrical load during a singlehalf-cycle of the AC power source.
 19. The load control device of claim18, wherein the control circuit is configured to determine azero-crossing time of a present half-cycle of the AC power source inresponse to a voltage developed across the thyristor.
 20. The loadcontrol device of claim 19, wherein the control circuit is configured tocontrol the gate coupling circuit to be conductive to conduct the gatecurrent through the gate terminal of the thyristor to render thethyristor conductive after a first, variable amount of time has elapsedsince the zero-crossing time of the present half-cycle of the AC powersource, and to maintain the thyristor conductive independent of themagnitude of the load current conducted through the thyristor during thepresent half-cycle, the control circuit configured to prevent the gatecoupling circuit from conducting the gate current after a second, fixedamount of time has elapsed since the zero-crossing time of the presenthalf-cycle to allow the thyristor to become non-conductive, the second,fixed amount of time being less than a length of the present half-cycle,the control circuit configured to vary the first, variable amount oftime in accordance with a determinate amount of power to be delivered tothe electrical load in order to cause the amount of power delivered tothe electrical load to be the determinate amount.
 21. The load controldevice of claim 20, wherein the control circuit comprises: a timingcircuit configured to generate a timing signal from the supply voltage,the timing circuit being configured to start to generate the timingsignal near the zero-crossing time of the present half-cycle, and tocause the timing signal to increase in magnitude with respect to time;and a drive circuit arranged to receive the timing signal, the drivecircuit being configured to render the thyristor conductive in eachhalf-cycle in response to the timing signal reaching a selectedthreshold; wherein the timing circuit is further configured to continuegenerating the timing signal after the thyristor is rendered conductivein each half-cycle, resulting in the control circuit maintaining thethyristor conductive independent of the magnitude of the load currentconducted through the thyristor, the timing circuit configured to stopgenerating the timing signal after the second, fixed amount of time haselapsed since the zero-crossing time of the present half-cycle, thedrive circuit configured to terminate the drive signal when the timingcircuit stops generating the timing signal so as to render the gatecoupling circuit non-conductive and prevent the gate terminal of thethyristor from conducting the gate current and thereby allow thethyristor to commutate off.
 22. The load control device of claim 18,wherein the control circuit comprises a microprocessor powered by thesupply voltage.
 23. The load control device of claim 18, furthercomprising: an RF receiver powered by the supply voltage and configuredto receive an RF signal; wherein the control circuit is configured todetermine the determinate amount of power to be delivered to theelectrical load from the RF signal.
 24. The load control device of claim18, wherein the electrical load is a high-efficiency lighting load andthe load control device comprises a dimmer switch.
 25. The load controldevice of claim 18, wherein the electrical load is a light-emittingdiode light source or a compact fluorescent lamp.
 26. The load controldevice of claim 18, wherein the power supply is configured to conductthe charging current through the electrical load when the thyristor isnon-conductive to generate the supply voltage for powering the controlcircuit.
 27. The load control device of claim 18, wherein the thyristorcomprises a triac.
 28. The load control device of claim 18, wherein thecontrol circuit is configured to generate the drive signal forcontrolling the gate coupling circuit to maintain the thyristorconductive independent of the magnitude of the load current conductedthrough the thyristor.